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| United States Patent |
5,835,529
|
|
Koga
, et al.
|
November 10, 1998
|
Digital communication apparatus
Abstract
A highly reliable and high-speed data transmission is made by using an FH
or MFSK mode as suitably selected according to the usage of channels. When
a reception signal is entered into a receiver through a transmission line,
(i) a signal processing unit supplies the spectrum intensity values of
carrier frequencies, (ii) a channel detection unit controls the phase of a
time slot based on the spectrum intensity values, selects the MFSK or FH
modulation mode, and supplies reception code data corresponding to
detected carrier frequencies, and (iii) a decoder supplies reception
information data based on the reception code data. When transmission
information data are entered into a transmitter, (i) a coding unit
supplies, according to the selected modulation mode, transmission code
data based on the transmission information data, (ii) a channel generation
unit supplies, based on the transmission code data, carrier frequencies to
be used and (iii) a waveform generation unit supplies a transmission
signal to a transmission line.
| Inventors:
|
Koga; Shoichi (Fukuoka, JP);
Igata; Yuji (Fukuoka, JP);
Hasako; Satoshi (Fukuoka, JP);
Maki; Masahiro (Fukuoka, JP);
Kishimoto; Michinori (Fukuoka, JP)
|
| Assignee:
|
Matsushita Electric Industrial Co., Ltd. (Osaka, JP)
|
| Appl. No.:
|
763338 |
| Filed:
|
December 11, 1996 |
Foreign Application Priority Data
| Dec 12, 1995[JP] | 7-322957 |
| Dec 12, 1995[JP] | 7-322967 |
| Jan 11, 1996[JP] | 8-002838 |
| Jul 12, 1996[JP] | 8-183000 |
| Nov 22, 1996[JP] | 8-311132 |
| Current U.S. Class: |
375/131; 370/344; 370/479; 375/134; 375/275; 375/303 |
| Intern'l Class: |
H04B 015/00 |
| Field of Search: |
375/202,324,303,200,275
370/344,479
|
References Cited [Referenced By]
U.S. Patent Documents
| 4271524 | Jun., 1981 | Goodman et al. | 375/202.
|
| 4320514 | Mar., 1982 | Haskell | 375/202.
|
| 4392231 | Jul., 1983 | Henry | 375/202.
|
| 5166953 | Nov., 1992 | Hershey et al. | 375/1.
|
| 5410538 | Apr., 1995 | Roche et al. | 370/479.
|
| 5442660 | Aug., 1995 | Kuo et al. | 375/202.
|
Other References
G. Einarsson, "Address Assignment for a Time-Frequency-Coded,
Spread-Spectrum System", Bell Systems Technical Journal, vol.59, No.7, pp.
1241-1255, Sep. 1980.
D. J. Goodman et al., "Frequency-Hopped Multilevel FSK for Mobile Radio",
Bell Systems Technical Journal, vol.59, No.7, pp. 1257-1275 Sep. 1980.
|
Primary Examiner: Chin; Stephen
Assistant Examiner: Liu; Shuwang
Attorney, Agent or Firm: McDermott, Will & Emery
Claims
What is claimed is:
1. In a digital communication apparatus to be used for a digital
communication system in which a plurality of digital communication
apparatus share a time slot and in which, using N carrier frequencies out
of M carrier frequencies per time slot, an N-channel frequency multiplex
communication is made with a multilevel frequency shift keying (MFSK)
modulation mode selected when N is equal to 1 and with a frequency hopping
(FH) modulation mode selected when N is not less than 2, each of N and M
being an integer, said digital communication apparatus comprising:
a receiver for supplying reception information data when a reception signal
is entered through a transmission line; and
a transmitter for supplying a transmission signal to said transmission line
when transmission information data are entered,
said receiver comprising:
a signal processing unit arranged such that, when said reception signal is
entered through said transmission line, there are calculated and supplied,
for said reception signal, the spectrum intensity values of said M carrier
frequencies per said time slot;
a channel detection unit arranged such that, based on said spectrum
intensity values, channels are detected, said time slot is controlled in
phase, either the MFSK or FH modulation mode is selected and reception
code data for said channels are supplied; and
a decoding unit for decoding said reception code data according to the
modulation mode thus selected, and for supplying said reception
information data, and
said transmitter comprising:
a coding unit arranged such that, when said transmission information data
are entered, said transmission information data are coded according to
said selected modulation mode and transmission code data are supplied;
a channel generation unit for assigning channels to said transmission code
data and for selecting and supplying carrier frequencies for said
channels; and
a waveform generation unit for supplying, as said transmission signal, the
signal waveforms of said selected carrier frequencies in synchronism with
said time slot to said transmission line.
2. A digital communication apparatus according to claim 1, wherein said
channel detection unit is arranged to detect, per time slot, the maximum
spectrum intensity value out of the spectrum intensity values of carrier
frequencies and to control said time slot in phase according to a
difference between the maximum spectrum intensity values at two
consecutive time slots.
3. A digital communication apparatus according to claim 1, wherein said
signal processing unit is arranged to intermittently execute a discrete
Fourier transform only at the time of the detection of carrier using a
carrier sense.
4. A digital communication apparatus according to claim 1, wherein said
coding unit is arranged to generate, at least one time, a transmission
code data string for upwardly frequency-sweeping (up-chirping) the carrier
frequencies from the least significant carrier frequency to the most
significant carrier frequency or for downwardly frequency-sweeping
(down-chirping) the carrier frequencies from the most significant carrier
frequency to the least significant carrier frequency, with a predetermined
period of time at the start of transmission set as a preamble.
5. A digital communication apparatus according to claim 4, wherein said
channel detection unit is arranged to calculate a frequency variable range
at the time of up-chirp detection or down-chirp detection, thereby to
digitally correct a frequency error in reference oscillating frequency
among a plurality of digital communication apparatus.
6. A digital communication apparatus according to claim 1, wherein said
channel generation unit is arranged to assign, according to a progressive
code, pieces of information to consecutive carrier frequencies.
7. In a digital communication apparatus to be used for a digital
communication system in which a plurality of digital communication
apparatus share a time slot and in which a frequency multiplex
communication is made with carrier frequencies out of M carrier
frequencies selected, per time slot, for a plurality of channels, M being
an integer not less than 2,
said digital communication apparatus comprising:
a transmitter for supplying a transmission signal to a transmission line
when transmission data are entered; and
a receiver for supplying reception data when a reception signal is entered
through said transmission line,
said transmitter comprising:
a frequency selection unit for determining, for said entered transmission
data, carrier frequencies to be used out of said M carrier frequencies per
log.sub.2 M bits according to a conversion table; and
a waveform generation unit for supplying, in synchronism with said time
slot, frequency waveforms corresponding to said carrier frequencies to be
used, said frequency waveforms being supplied, as said transmission
signal, to said transmission line per period of one time slot T,
said receiver comprising:
a down-converter unit for down-converting in frequency a reception signal
entered through said transmission line to a low frequency band;
a DFT operation unit for successively executing, per sampling clock period
.DELTA.t, a discrete Fourier transform (DFT) for a period of the latest
one time slot (T=N.times..DELTA.t) on said signal after down-converted in
frequency, thereby to calculate spectrum values I (k) (k=1, 2, . . . , M)
respectively for said M carrier frequencies, N being an integer not less
than M;
a threshold judgment unit for detecting, out of said M carrier frequencies,
carrier frequencies of which spectrum values I(k) exceed a threshold
value, said carrier frequencies being detected as candidate carrier
frequencies per said sampling clock cycle .DELTA.t;
a synchronizing signal generation unit for generating, based on said
spectrum values I(k) and said candidate carrier frequencies, a
synchronizing trigger signal for synchronization with said time slot;
a latch unit for determining, as reception carrier frequencies, said
candidate carrier frequencies at the time of assertion of said
synchronizing trigger signal; and
a decoder for supplying, based on a conversion table identical with that in
said frequency selection unit, log.sub.2 M-bit reception data for each of
said reception carrier frequencies.
8. A digital communication apparatus according to claim 7, wherein said
waveform generation unit is arranged such that,
using i) cosine waves represented by cos
(2.pi..times..DELTA.f.times.(2k-1).times.t), ii) sine waves represented by
(-1).sup.k-1 .times.sin (2.pi..times..DELTA.f.times.(2k-1).times.t) and
iii) two carriers having frequency fc and different in phase respectively
represented by cos (2.pi..times.fc.times.t) and sin
(2.pi..times.fc.times.t), in which .DELTA.f is the frequency step width
equal to 1/T.times.R and t is time, R is an integer not less than 1;
frequency waveforms W1 which are represented by the following equation:
W1=sin (2.pi..times.(fc+(-1).sup.k-1 .times..DELTA.f.times.(2k-1).times.t)
and which are corresponding to said carrier frequencies to be used, are
supplied, per period of one time slot T, by a frequency orthogonal
transformation in synchronism with said time slot, said frequency
waveforms W1 being supplied as said transmission signal to said
transmission line, and
said down-converter unit is arranged such that, using said frequency fc, a
reception signal entered through said transmission line is down-converted
in frequency to a low frequency band.
9. A digital communication apparatus according to claim 7, wherein said
waveform generation unit is arranged such that,
using i) cosine waves represented by cos
(2.pi..times..DELTA.f.times.(2k-1).times.t), ii) sine waves represented by
(-1).sup.k .times.sin (2.pi..times..DELTA.f.times.(2k-1).times.t) and iii)
two carriers having frequency fc and different in phase respectively
represented by cos (2.pi..times.fc.times.t) and sin
(2.pi..times.fc.times.t), in which .DELTA.f is the frequency step width
which is equal to 1/T.times.R and t is time, R is an integer not less than
1;
frequency waveforms W2 which are represented by the following equation:
W2=sin (2.pi..times.(fc+(-1).sup.k .times..DELTA.f.times.(2k-1).times.t)
and which are corresponding to said carrier frequencies to be used, are
supplied, per period of one time slot T, by a frequency orthogonal
transformation in synchronism with said time slot, said frequency
waveforms W2 being supplied as said transmission signal to said
transmission line, and
said down-converter unit is arranged such that, using said frequency fc, a
reception signal entered through said transmission line band is
down-converted in frequency to a low frequency band.
10. A digital communication apparatus according to claim 7, wherein said
waveform generation unit is arranged such that,
using i) cosine waves represented by cos
(2.pi..times..DELTA.f.times.k.times.t), ii) sine waves represented by
(-1).sup.k-1 .times.sin (2.pi..times..DELTA.f.times.k.times.t) and iii)
two carriers having frequency fc and different in phase respectively
represented by cos (2.pi..times.fc.times.t) and sin
(2.pi..times.fc.times.t), in which .DELTA.f is the frequency step width
which is equal to 1/T.times.R and t is time, R is an integer not less than
1;
frequency waveforms W3 which are represented by the following equation:
W3=sin (2.pi..times.(fc+(-1).sup.k-1 .times..DELTA.f.times.k.times.t)
and which are corresponding to said carrier frequencies to be used, are
supplied, per period of one time slot T, by a frequency orthogonal
transformation in synchronism with said time slot, said frequency
waveforms W3 being supplied as said transmission signal to said
transmission line, and
said down-converter unit is arranged such that, using said frequency fc, a
reception signal entered through said transmission line is down-converted
in frequency to a low frequency band.
11. A digital communication apparatus according to claim 7, wherein said
waveform generation unit is arranged such that,
using i) cosine waves represented by cos
(2.pi..times..DELTA.f.times.k.times.t), ii) sine waves represented by
(-1).sup.k .times.sin (2.pi..times..DELTA.f.times.k.times.t) and iii) two
carriers having frequency fc and different in phase respectively
represented by cos (2.pi..times.fc.times.t) and sin
(2.pi..times.fc.times.t), in which .DELTA.f is the frequency step width
which is equal to 1/T.times.R and t is time, R is an integer not less than
1;
frequency waveforms W4 which are represented by the following equation:
W4=sin (2.pi..times.(fc+(-1).sup.k .times..DELTA.f.times.k.times.t)
and which are corresponding to said carrier frequencies to be used, are
supplied, per period of one time slot T, by a frequency orthogonal
transformation in synchronism with said time slot, said frequency
waveforms W4 being supplied as said transmission signal to said
transmission line, and
said down-converter unit is arranged such that, using said frequency fc, a
reception signal entered through said transmission line is down-converted
in frequency to a low frequency band.
12. A digital communication apparatus according to claim 7, wherein said
DFT operation unit is arranged to execute said discrete Fourier transform
(DFT) using, as sampling frequency (1/.DELTA.t), frequency
(M.times.4.times..DELTA.t) equal to the occupied frequency bandwidth of
said M carrier frequencies.
13. A digital communication apparatus according to claim 7, wherein there
is determined, in said conversion table in said frequency selection unit,
the relationship between the arrangement of said carrier frequencies after
down-converted in frequency and said transmission data, said relationship
being determined based on a progressive code.
14. A digital communication apparatus according to claim 7, wherein said
synchronizing signal generation unit is arranged:
to calculate, per sampling clock cycle .DELTA.t, a cost function Cf defined
by the following equation:
Cf=.SIGMA..sub.k=1.sup.M (I(k))-.SIGMA..sub.l=0.sup.s (Id(l))
in which s is an integer not less than 1 and not greater than M and Id(l)
(l=0, . . . , s) is the spectrum value of each of said candidate carrier
frequencies;
to hold, per time point i (i=1, 2, . . . , N) in steps of .DELTA.t of said
time slot, the accumulated value C(i) of the cost functions Cf for the
latest TC time slots determined by TC, in which TC is time constant and is
an integer not less than 1; and
to detect, per said time slot, the time point at which the smallest
accumulated value C(i) is held, thereby to generate said synchronizing
trigger signal.
15. A digital communication apparatus according to claim 14, wherein said
transmission data entered into said frequency selection unit have a
randomized property.
16. A digital communication apparatus according to claim 14, wherein said
synchronizing signal generation unit is arranged to generate said
synchronizing trigger signal with said time constant TC set to M.times.q
in which q is an integer not less than 1.
17. A digital communication apparatus according to claim 7, wherein:
said down-converter unit further has a function of normalizing, using an
automatic gain control (AGC) amplifier, the amplitude of said signal after
down-converted in frequency; and
said threshold judgment unit is arranged to set said threshold value
according to the largest spectrum value out of the spectrum values of said
candidate carrier frequencies.
18. A digital communication apparatus according to claim 7, further
comprising a frequency control unit for finely adjusting, according to a
difference in spectrum value between certain carrier frequency and carrier
frequency adjacent thereto, reference frequency fc used in said waveform
generation unit and said down-converter unit.
Description
BACKGROUND OF THE INVENTION
The present invention relates to a digital communication apparatus.
In the communication field, a spread-spectrum communication technique is
suitable for a high-speed data transmission in the environment where the
channel characteristics such as multipath fading undergo a considerable
dynamic change.
Typical examples of spread-spectrum communication technique include a
direct spread (DS) system and a frequency hopping (FH) system. The DS
system is advantageous in view of small circuit size and high-speed data
transmission, while the FH system is advantageous in view of channel
capacity and communication reliability. Examples of the FH system include
a high-speed FH system and a low-speed FH system. The high-speed FH system
in which communication is made while the carrier frequency is being
switched in a short period of time, is considerably increased in hardware
size as compared with the low-speed FH system, but is advantageous in view
of reliability against multipath fading.
Examples of a primary modulation in the FH system include a frequency shift
keying (FSK) modulation, a phase shift keying (PSK) modulation and the
like. In view of simplicity in circuit configuration requiring no phase
control, the FSK modulation is relatively often used.
According to an arrangement-of an FH digital communication apparatus of
prior art, the transmission throughput per channel, even for one-channel
communication, is the same as that in communication using a plurality of
channels.
According to another arrangement of the FH digital communication apparatus
of prior art, carrier frequency waveforms are synthesized by a PLL
synthesizer in the transmitter. This makes it difficult to switch the
carrier frequency at a high speed of the order of micro second. Thus, such
an arrangement is not suitable for the high-speed FH system. Further, the
receiver requires, at its envelop line detector unit, analog band-pass
filters having sharp amplitude characteristics in number equal to the
number of carrier frequencies. This results in an increase in hardware. To
achieve the high-speed FH system, it would be proposed that both the
generation of waveforms and the detection of frequencies are conducted by
a digital signal process. However, this disadvantageously excessively
increases the frequency of a sampling clock for a digital signal process.
On the other hand, when detecting frequencies using a discrete Fourier
transform (DFT), it is required that the DFT operation interval is
accurately in synchronism with the time slot. This has hitherto been
difficult.
There is known a digital communication apparatus using a code multiplexing
MFSK modulation using M carrier frequencies, M being an integer not less
than 4. According to D. J. Goodman et al., "Frequency-Hopped Multilevel
FSK for Mobile Radio", Bell System Technical Journal, Vol. 59, No. 7, pp.
1257-1275, September 1980, M frequencies (tones) are prepared in a
predetermined band according to the high-speed FH system, and a unique
code is assigned to each user on a time-frequency matrix. However, a high
sampling rate is required in the DFT process, making it practically
difficult to achieve the hardware.
There is now considered a digital communication apparatus of the mode
changeover type arranged to make a frequency multiplex communication with
either the MFSK or FH mode selected according to multiplicity. However,
when the transmitter is not provided with a data scrambling function and
the appearance probability of transmission data is uneven, the spectra of
a transmission signal are also uneven. Further, when specific frequency
components appear continuously, timing extraction becomes difficult in the
receiver. This lengthens the time required for pulling into synchronism.
Further, in the receiver, there are instances where, in an operation mode
according to the MFSK mode, a plurality of reception signals are detected
under the influence of noise, a spurious response or the like. In such a
case, the maximum likelihood word cannot be determined. Further, in an
operation mode according to the FH mode, too, when a plurality of words
are calculated by a majority judgment, the maximum likelihood word can
neither be determined.
In G. Einarsson, "Address Assignment for a Time-Frequency-Coded,
Spread-Spectrum System", Bell System Technical Journal, Vol. 59, No. 7, pp
1241-1255, September 1980, two methods are proposed for generating hopping
codes from data in a digital FH-MFSK communication system. One is based on
the premise of a synchronous system, while the other is based on the
premise of an asynchronous system (a code multiplexing system providing a
chip synchronism between users, but not providing a frame synchronism
between users). Both methods are based on a Reed-Solomon code. However,
under the influence of frequency-selective fading, there might occur a
miss detection (deletion) of all specific frequency components.
SUMMARY OF THE INVENTION
It is an object of the present invention to provide a digital communication
apparatus in which a high-speed data transmission is made using a
multilevel frequency shift keying (MFSK) modulation mode when all the
channels become vacant.
It is another object of the present invention to provide a digital
communication apparatus in which a data communication is made according to
a high-speed FH mode with no considerable increase in both sampling clock
frequency and hardware size even though reception carrier frequencies are
detected by a DFT operation unit in the receiver.
It is a further object of the present invention to provide a digital
communication apparatus in which, using a low sampling-rate DFT processor
capable of processing a 1/2 band width of a sub-band, a specific sub-band
is modulated/demodulated according to the MFSK or code multiplexing MFSK
mode even in the environment where simultaneous communications are made
using a plurality of sub-bands.
It is still another object of the present invention to provide a digital
communication apparatus of the mode changeover type capable of randomizing
transmission data without use of a scrambler and having maximum likelihood
word determining means.
It is a still further object of the present invention to provide a digital
communication apparatus highly invulnerable to fading such that random
hopping codes are acquired.
To achieve the objects above-mentioned, the present invention provides a
digital communication apparatus to be used for a communication system in
which a plurality of digital communication apparatus share a time slot
(network synchronization) and in which, using N carrier frequencies out of
M carrier frequencies per time slot, an N-channel frequency multiplex
communication is made with an MFSK modulation mode selected when N is
equal to 1 and with an FH modulation mode selected when N is not less than
2, each of N and M being an integer. More specifically, the digital
communication apparatus of the present invention comprises: the following
receiver comprising a signal processing unit, a channel detection unit and
a decoding unit; and the following transmitter comprising a coding unit, a
channel generation unit and a waveform generation unit. In the receiver,
the signal processing unit is arranged such that, when a reception signal
is entered through a transmission line, there are -calculated, for the
reception signal, the spectrum intensity values of the M carrier
frequencies per time slot, and that the spectrum intensity values thus
calculated are supplied to the channel detection unit. The channel
detection unit is arranged such that, when the spectrum intensity values
are entered from the signal processing unit, channels are detected based
on the spectrum intensity values, that the time slot is controlled in
phase, that either the MFSK or FH modulation mode is selected and that
reception code data for the channels are supplied to the decoding unit.
The decoding unit is arranged such that, when reception code data are
entered from the channel detection unit, the reception code data are
decoded according to the modulation mode selected by the channel detection
unit, and that reception information data are supplied. In the
transmitter, the coding unit is arranged such that, when transmission
information data are entered, the transmission information data are coded
according to the modulation mode selected by the channel detection unit
and that transmission code data are supplied to the channel generation
unit. The channel generation unit is arranged to assign channels to the
transmission code data received from the coding unit, to select carrier
frequencies for the channels and to supply the carrier frequencies thus
selected to the waveform generation unit. The waveform generation unit is
arranged to supply, as a transmission signal, the signal waveforms of the
carrier frequencies selected by the channel generation unit, the
transmission signal being supplied, in synchronism with the time slot, to
the transmission line. According to the digital communication apparatus
having the arrangement above-mentioned, the modulation mode can be
switched from the FH mode to the MFSK mode and vice versa merely by
changing the contents to be processed in the coding and decoding units.
This enables the FH or MFSK mode to be used as properly selected according
to the usage of channels. This achieves an efficient high-speed data
transmission without reliability lost.
In a digital communication system using another digital communication
apparatus according to the present invention, a plurality of digital
communication apparatus share a time slot (network synchronization) and a
frequency multiplex communication is made with carrier frequencies out of
M carrier frequencies selected, per time slot, for a plurality of
channels, M being an integer not less than 2. This digital communication
apparatus comprises; the following transmitter comprising a frequency
selection unit and a waveform generation unit; and the following receiver
comprising a down-converter unit, a DFT operation unit, a threshold
judgment unit, a synchronizing signal generation unit, a latch unit and a
decoder. In the transmitter, the frequency selection unit is arranged to
determine, for entered transmission data, carrier frequencies to be used
out of the M carrier frequencies per log.sub.2 M bits according to a
conversion table. The waveform generation unit is arranged to supply, in
synchronism with the time slot, frequency waveforms corresponding to the
carrier frequencies to be used, the frequency waveforms being supplied, as
a transmission signal, to the transmission line for each period of one
time slot T. In the receiver, the down-converter unit is arranged such
that a reception signal entered through the transmission line is
down-converted in frequency to a low frequency band. The DFT operation
unit is arranged to successively execute, per sampling clock period At, a
discrete Fourier transform (DFT) for a period of the latest one time slot
(T=N.times..DELTA.t) on the signal after down-converted in frequency,
thereby to respectively calculate spectrum values I (k) (k=1, 2, . . . ,
M) for the M carrier frequencies, N being an integer not less than M. The
threshold judgment unit is arranged to detect, out of the M carrier
frequencies, carrier frequencies of which spectrum values I(k) exceed a
threshold value, these carrier frequencies being detected as candidate
carrier frequencies per sampling clock period .DELTA.t. The synchronizing
signal generation unit is arranged to generate, based on the spectrum
values I(k) and the candidate carrier frequencies, a synchronizing trigger
signal for synchronization with the time slot. The latch unit is arranged
to determine, as reception carrier frequencies, the candidate carrier
frequencies at the time of assertion of the synchronizing trigger signal.
The decoder is arranged to supply, based on a conversion table identical
with that in the frequency selection unit, log.sub.2 M-bit reception data
for the reception carrier frequencies. According to the digital
communication apparatus having the arrangement above-mentioned, a
transmission signal can be pulled, using the results of a DFT operation,
into accurate synchronism with the time slot. Thus, such a highly precise
frequency detection suitable for the high-speed FH system achieves a
highly reliable data communication with a high frequency-utilization
efficiency.
The present invention provides a further digital communication apparatus
using either an MFSK modulation mode or a code multiplexing MFSK
modulation mode, using M carrier frequencies per sub-band, M being an
integer not less than 4, and this digital communication apparatus is
arranged such that the M carrier frequencies per sub-band are orthogonally
disposed at frequency intervals not less than 2/T in which T is a
frequency switching period of time. According to the digital communication
apparatus having the arrangement above-mentioned, using a low
sampling-rate discrete Fourier transform capable of processing a 1/2 band
width of a sub-band, frequencies in the sub-band around a specific
frequency can be detected even in the environment where simultaneous
communications are made using a plurality of sub-bands.
The present invention provides a further digital communication apparatus
using either an MFSK modulation mode or a code multiplexing MFSK
modulation mode, using M consecutive carrier frequencies randomly selected
per predetermined time interval L, M being an integer not less than 4, and
this digital communication apparatus is arranged such that the time
interval L is a value equal to the product of a frequency switching period
of time T and a positive integer and that the M carrier frequencies are
orthogonally disposed at frequency intervals not less than 2/T. According
to the digital communication apparatus having the arrangement
above-mentioned, using a low sampling-rate discrete Fourier transform
capable of processing a 1/2 band width of a sub-band, frequencies in the
sub-band around the desired frequency can be detected even in the
environment where simultaneous communications are made using a plurality
of sub-bands.
The present invention provides a further digital communication apparatus
using either an MFSK modulation mode or a code multiplexing MFSK
modulation mode, using M carrier frequencies per sub-band, M being an
integer not less than 4, and this digital communication apparatus
comprises a transmitter and a receiver. The receiver comprises: N
diversity branches in which signals received from N points spatially
separated from the diversity branches, are respectively down-converted in
frequency to low frequency bands, thereby to supply N-sequence base band
signals, N being an integer not less than 2; a frequency detection unit
formed of M operation units for respectively calculating the signal levels
of the M carrier frequencies; a selector for assigning the N-sequence base
band signals to the M operation units; and a timer for controlling the
selector to change the base band signal to be assigned to a specific
operation unit out of the M operation units when the signal level
calculated by the specific operation unit does not exceed a threshold
level in a predetermined period of time. According to the digital
communication apparatus having the arrangement above-mentioned, signal
reception can be made with no fading influence in each of the operation
units.
The present invention provides a further digital communication apparatus to
be used for a digital communication system in which a plurality of digital
communication apparatus share a time slot and in which a half-duplex data
communication is made using either an MFSK modulation mode or a code
multiplexing MFSK modulation mode, and this digital communication
apparatus comprises a transmitter and a receiver which share a single
antenna. In this digital communication apparatus, the receiver comprises:
first means for storing, as a reference phase error, a phase error which
is present immediately before the communication mode is switched from the
reception mode to the transmission mode; and second means for generating,
after the reception mode has been switched to the transmission mode, a
regenerative synchronizing signal for synchronous control of the time
slot, using a feedforward control based on the stored reference phase
error, and for supplying the regenerative synchronizing signal thus
generated to the transmitter. According to the digital communication
apparatus having the arrangement above-mentioned, it is possible to
maintain a network synchronization at the time when there is made, using
the common antenna, a code division multiple access (CDMA) as done in an
FH-MFSK mode in the same frequency band.
The present invention provides a further digital communication apparatus
comprising: a transmitter in which a convolutional coder and an
interleaver are combined to code transmission data without use of a
scrambler; and a receiver in which a majority decoder is used to execute a
most likelihood word decoding. In this digital communication apparatus,
using M carrier frequencies per time slot, a frequency multiplex
communication is made with either an MFSK modulation mode or an FH
modulation mode selected according to multiplicity, M being an integer not
less than 2. This digital communication apparatus comprises (i) the
transmitter comprising: the convolutional coder for supplying a
convolutional code sequence according to an input information sequence;
the interleaver for supplying an interleave sequence according to the
convolutional code sequence; an FH coder for supplying an FH code sequence
according to the interleave sequence; a first switching unit for
supplying, according to a switching signal, either the interleave sequence
or the FH code sequence as a transmission sequence; and an M-ary
independent signal transmitter unit for supplying, per time slot, a
transmission signal containing, out of M mutually independent frequency
components, one frequency component corresponding to the transmission
sequence, and (ii) the receiver comprising: an M-ary independent signal
receiver unit for supplying a threshold judgment pattern generated by
making a threshold judgment on each of the intensity values of M frequency
components of a reception signal; an operational mode control circuit for
judging the multiplicity based on the threshold judgment pattern and for
supplying the switching sinal according to the multiplicity; an FH decoder
for supplying an FH decoding pattern according to the threshold judgment
pattern; a second switching unit for selecting, according to the switching
signal, either the threshold judgment pattern or the FH decoding pattern;
the majority decoder for supplying a majority decoding sequence according
to the pattern selected by the second switching unit; a deinterleaver for
supplying a deinterleave sequence according to the majority decoding
sequence; and a Viterbi decoder for supplying an information sequence
according to the deinterleave sequence. According to the digital
communication apparatus having the arrangement above-mentioned, both the
convolutional coder and the interleaver encode transmission data, causing
the transmission data to be randomized without use of a scrambler. This
not only equalizes the spectra of a transmission signal, but also reduces
the frequency in continuous appearance of specific frequency components.
Further, the majority decoder in the receiver makes a majority judgment on
each of the bits forming a word, thus determining the most likelihood
word.
The present invention provides a further digital communication apparatus
comprising a frequency hopping generator (FH coder) comprising the
following conversion means and the following operation means. More
specifically, the conversion means is arranged to convert a data value x
which is an element of a Galois field, into a code w which is a non-zero
element of the Galois field, according to the following conversion
equation using a function f:
w=f(x)
when the number M of values which a data can present, is equal to 2.sup.k
(k is a positive integer) and the number Q of the elements of the Galois
field is equal to p.sup.r (>M) in which p is a prime number and r is a
positive integer. The operation means is to arrange to calculate,
according to the code w, a hopping code vector y composed of L components
using the following Galois operation:
y=w.times. .alpha.+i.multidot. e
wherein i is the user identification No. which is an element of the Galois
field; .alpha. is one of the primitive elements of the Galois field;
.alpha. is a spread code vector of L components and is equal to (1,
.alpha., .alpha..sup.2, . . . .alpha..sup.L-1) in which L is an integer
not less than 2 and not greater than p.sup.r -1; and e is a unit vector
of L components and is equal to (1, 1, . . . , 1). According to the
digital communication apparatus having the arrangement above-mentioned, Q
is greater than M and the data value x is previously converted into the
non-zero code w, based on which the hopping code vector y is calculated.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram showing an example of the arrangement of a
digital communication apparatus according to the present invention;
FIG. 2 is a block diagram showing in detail an example of the arrangement
of the transmitter in FIG. 1;
FIG. 3 is a block diagram showing in detail an example of the arrangement
of the receiver in FIG. 1;
FIG. 4 is a block diagram showing in detail an example of the arrangement
of the coding unit in the transmitter in FIG. 2;
FIG. 5 shows the contents of the frequency table in the transmitter in FIG.
2:
FIG. 6 shows the contents of the frequency table in the receiver in FIG. 3:
FIG. 7 shows the operational timings at the time when the FH mode is
selected in the circuit in FIG. 4;
FIG. 8 shows the operational timings at the time when the MFSK mode is
selected in the circuit in FIG. 4;
FIG. 9 shows the waveform of a digital synthesizer output together with the
waveform of a time slot signal in the case in FIG. 8;
FIG. 10 is a view of transition of carrier frequencies to be used in the
case in FIG. 8;
FIG. 11 is an input timing diagram of a DFT process in a phase asynchronous
state;
FIG. 12 is an input timing diagram of a DFT process in a phase synchronous
state;
FIG. 13 shows the state of carrier frequencies to be used when the MFSK
mode is selected:
FIG. 14 shows the state of carrier frequencies to be used when the FH mode
is selected:
FIG. 15 is a block diagram showing in detail an example of the arrangement
of the time slot signal generation unit in the channel detection unit in
FIG. 3;
FIG. 16 is a block diagram showing a modification of the signal processing
unit in FIG. 3;
FIG. 17 is a block diagram showing a modification of the coding unit in
FIG. 2;
FIG. 18 shows up-chirp code data in the coding unit in FIG. 17;
FIG. 19 shows down-chirp code data in the coding unit in FIG. 17;
FIG. 20 is a block diagram showing a modification of the receiver in FIG.
3, corresponding to FIG. 17;
FIG. 21 is a view of frequency transition of a chirp signal in the
arrangement in FIGS. 17 and 20;
FIG. 22 is a view illustrating a DFT process on the least significant
carrier frequency in the arrangements in FIGS. 17 and 20:
FIG. 23 is a view illustrating a DFT process on the most significant
carrier frequency in the arrangements in FIGS. 17 and 20:
FIG. 24 is a view of a modification of the arrangement in FIG. 5;
FIG. 25 is a view of a modification of the arrangement in FIG. 6;
FIG. 26 is a view of a modification of the arrangement in FIG. 18;
FIG. 27 is a view of a modification of the arrangement in FIG. 19;
FIG. 28 is a block diagram showing an example of the arrangement of the
digital communication apparatus according to the present invention;
FIG. 29 shows a correspondence of 4-bit data to carrier frequencies in the
frequency selection unit and the decoder in FIG. 28;
FIG. 30 is a block diagram showing in detail an example of the arrangement
of the cosine-wave and sine-wave generation unit in FIG. 28;
FIG. 31 shows an example of the arrangement of carrier frequencies after
frequency orthogonal transformation in the waveform generation unit in
FIG. 28;
FIG. 32 shows another example of the arrangement of carrier frequencies
after frequency orthogonal transformation;
FIG. 33 shows an example of the arrangement of carrier frequencies after
down-conversion, corresponding to the arrangement in FIG. 31;
FIG. 34 shows a further example of the arrangement of carrier frequencies
after frequency orthogonal transformation;
FIG. 35 shows a further example of the arrangement of carrier frequencies
after frequency orthogonal transformation;
FIG. 36 is a block diagram showing in detail a circuit arrangement for a
DFT operation for one carrier frequency in the DFT operation unit in FIG.
28;
FIG. 37 is a block diagram showing in detail an example of the arrangement
of the threshold judgment unit in FIG. 28;
FIG. 38 is a block diagram showing in detail an example of the arrangement
of the threshold value control unit in FIG. 37;
FIG. 39 is a block diagram showing in detail an example of the arrangement
of the synchronizing signal generation unit in FIG. 28;
FIG. 40 is a block diagram showing in detail an example of the arrangement
of the clock regeneration unit in FIG. 39;
FIG. 41 is a diagram of operational timing of the clock regeneration unit
in FIG. 40;
FIG. 42 shows two reception channels high in randomized property;
FIG. 43 shows an example of the waveform of a cost function accumulated
value in the synchronizing signal generation unit in FIG. 39 when two
channels in FIG. 42 are received;
FIG. 44 shows examples of two reception channels low in randomized
property;
FIG. 45 shows an example of the waveform of a cost function accumulated
value in the synchronizing signal generation unit in FIG. 39 when two
channels in FIG. 44 are received;
FIG. 46 shows examples of three reception channels high in randomized
property;
FIG. 47 shows an example of the waveform of a cost function accumulated
value in the synchronizing signal generation unit in FIG. 39 when three
channels in FIG. 46 are received;
FIG. 48 shows examples of three reception channels low in randomized
property;
FIG. 49 shows an example of the waveform of a cost function accumulated
value in the synchronizing signal generation unit in FIG. 39 when three
channels in FIG. 48 are received;
FIG. 50 shows examples of one reception channel;
FIG. 51 shows an example of the waveform of a cost function accumulated
value in the synchronizing signal generation unit in FIG. 39 when a
channel in FIG. 50 is received under a noise environment with time
constant TC being equal to 1;
FIG. 52 shows an example of the waveform of a cost function accumulated
value in the synchronizing signal generation unit in FIG. 39 when a
channel in FIG. 50 is received under a noise environment with time
constant TC being equal to 16;
FIG. 53 is a block diagram illustrating a modification of the digital
communication apparatus in FIG. 28;
FIG. 54 shows the levels of spurious responses which reception carrier
frequency gives to adjacent frequency bands in a comparative example of
the digital communication apparatus in FIG. 53;
FIG. 55 is a block diagram illustrating an example of the arrangement of
the digital communication apparatus according to the present invention;
FIG. 56 is a diagram of frequency arrangement, at a certain time, of three
sub-bands used in the digital communication apparatus in FIG. 55;
FIG. 57 is a diagram of frequency arrangement obtained after
down-conversion of a first sub-band;
FIG. 58 is a diagram of frequency arrangement obtained after
down-conversion of a second sub-band;
FIG. 59 is a diagram of frequency arrangement obtained after
down-conversion of a third sub-band;
FIG. 60 is a block diagram illustrating a modification of the arrangement
in FIG. 55;
FIG. 61 is a diagram of frequency arrangement, at a certain time, of two
sub-bands used in the digital communication apparatus in FIG. 60;
FIG. 62 is a diagram of frequency arrangement obtained after
down-conversion of a first sub-band;
FIG. 63 is a diagram of frequency arrangement obtained after
down-conversion of a second sub-band;
FIG. 64 is a block diagram illustrating a modification of the receiver in
FIG. 55;
FIG. 65 is a block diagram illustrating in detail an example of the
arrangement of one operation unit in FIG. 64;
FIGS. 66A, 66B and 66C show frequencies received under the influence of
fading in the diversity branches in FIG. 64;
FIG. 67 is a block diagram illustrating a modification of the arrangement
in FIG. 55;
FIG. 68 is a block diagram illustrating in detail an example of the
arrangement of the window control unit in FIG. 67;
FIG. 69 is a diagram of operational timing of the window control unit in
FIG. 68;
FIG. 70 is a block diagram illustrating an example of the arrangement of
the digital communication apparatus according to the present invention;
FIG. 71 is a block diagram showing in detail an arrangement of the
convolutional coder in FIG. 70;
FIGS. 72A and 72B are block diagrams respectively illustrating in detail
the arrangements of the interleaver and the deinterleaver in FIG. 70;
FIGS. 73A and 73B are block diagrams respectively illustrating in detail
the arrangements of the FH coder and the FH decoder in FIG. 70;
FIGS. 74A, 74B and 74C respectively show examples of an interleave sequence
matrix, a multiplexing code matrix and an FH code sequence matrix in the
FH coder in FIG. 73A;
FIGS. 75A, 75B, 75C and 75D respectively show examples of a threshold
judgment pattern, a multiplexing code sequence, a judgment matrix and an
FH decoding pattern matrix in the FH decoder in FIG. 73B;
FIGS. 76A and 76B are block diagrams respectively illustrating in detail
the arrangements of the M-ary independent signal transmission unit and the
M-ary independent signal reception unit in FIG. 70;
FIG. 77 is a block diagram illustrating in detail an example of the
arrangement of the operational mode control circuit in FIG. 70;
FIG. 78 is a block diagram illustrating in detail an example of the
arrangement of the majority decoder in FIG. 70;
FIG. 79 is a block diagram illustrating a modification of the arrangement
in FIG. 70;
FIG. 80 is a block diagram illustrating in detail an example of the
arrangement of the burst signal component: removal circuit in FIG. 79;
FIG. 81 is a block diagram illustrating in detail an example of the
arrangement of each of 16 burst detection units forming the burst
detection circuit in FIG. 80;
FIG. 82 is a block diagram illustrating in detail an example of the
arrangement of the burst removal circuit in FIG. 80;
FIG. 83 is a block diagram illustrating in detail an example of the
arrangement of each of 16 burst removal logic units in FIG. 82;
FIG. 84 is a block diagram illustrating a further modification of the
arrangement in FIG. 70;
FIG. 85 is a block diagram illustrating in detail an example of the
arrangement of the puncture signal generator in FIG. 84;
FIG. 86 is a block diagram illustrating a further modification of the
arrangement in FIG. 70;
FIG. 87 is a block diagram illustrating in detail an example of the
arrangement of the multi-level decoder in FIG. 86;
FIGS. 88A and 88B are block diagrams respectively illustrating
modifications of the arrangements in FIGS. 76A, 76B;
FIG. 89 is a block diagram illustrating a further modification of the
arrangement in FIG. 70;
FIGS. 90A and 90B are block diagrams respectively illustrating in detail
the arrangements of the FH coder and the FH decoder in FIG. 89;
FIG. 91 is a block diagram illustrating in detail an example of the
arrangement of the operational mode control circuit in FIG. 89;
FIG. 92 shows the relationship between input and output of the multiplicity
judgment logic in FIG. 91;
FIG. 93 is a block diagram illustrating an FH-MFSK digital communication
system of prior art;
FIGS. 94A and 94B respectively illustrate the definitions of Galois
addition and Galois multiplication used in the FH code generator in FIG.
93;
FIG. 95 shows examples of a hopping code vector generated in the FH code
generator in FIG. 93;
FIGS. 96A and 96B show time/frequency matrices, under the influence of
frequency-selective fading, in the transmitter and receiver in FIG. 93;
FIGS. 97A and 97B respectively illustrate the definitions of Galois
addition and Galois multiplication used in the FH code generator in the
digital communication system according to the present invention;
FIG. 98 shows examples of a hopping code vector in the digital
communication system according to the present invention;
FIGS. 99A and 99B show time/frequency matrices, under the influence of
frequency-selective fading, in the transmitter and receiver in the digital
communication system according to the present invention;
FIG. 100 is a block diagram showing the arrangement of the FH code
generator in the digital communication system according to the present
invention;
FIG. 101 shows the operation of the chip counter in FIG. 100;
FIG. 102 shows the operation of the data conversion unit in FIG. 100;
FIG. 103 shows the operation of the spread code generator in FIG. 100;
FIG. 104 shows a modification of the arrangement in FIG. 100;
FIG. 105 shows a further modification of the arrangement in FIG. 100;
FIG. 106 shows the operation of the FH code judgment unit in FIG. 105; and
FIG. 107 shows the relationship between the number M of values which a data
can present and the number Q of the elements of a Galois field, in the FH
code generator in the digital communication system according to the
present invention.
DETAILED DESCRIPTION OF THE INVENTION
The following description will discuss embodiments of a digital
communication apparatus according to the present invention with reference
to the attached drawings.
FIG. 1 shows an example of the arrangement of a digital communication
apparatus according to the present invention. In FIG. 1, the digital
communication apparatus comprises a transmitter T, a receiver R, a coding
unit CO, a channel generation unit CG, a waveform generation unit WG, a
signal processing unit DS, a channel detection unit CD and a decoding unit
DE. The transmitter T comprises the coding unit CO, the channel generation
unit CG and the waveform generation unit WG. The receiver R comprises the
signal processing unit DS, the channel detection unit CD and the decoding
unit DE. Shown in FIG. 1 are transmission information data dt,
transmission code data coo, carrier frequencies to be used cgo, a
transmission signal ot, a reception signal or, a spectrum intensity output
dso, reception code data cdo, reception information data dr, a time slot
signal ts and a mode control signal mo.
In FIG. 1, the binary transmission information data dt are entered into the
transmitter T. In the coding unit CO, the transmission information data dt
are subjected to error correction coding and then, based on the state of
the mode control signal mo, the transmission code data coo according to
the FH or MFSK mode are generated. In the channel generation unit CG,
based on the block of log.sub.2 M bits of the transmission code data coo,
the carrier frequencies to be used cgo are read out from a memory table
containing pieces of carrier frequency information respectively assigned
to the blocks. Here, M is an integer and represents the maximum number of
carrier frequencies to be utilized in the frequency multiple communication
system. In the waveform generation unit WG, the signal waveforms of the
carrier frequencies to be used cgo for one symbol, are digitally generated
by a digital direct synthesizer (hereinafter simply referred to as digital
synthesizer) and then supplied as the transmission signal ot to a
transmission line in synchronism with the time slot signal ts for setting
the symbol interval.
The receiver R receives the reception signal or from a transmission line.
In the signal processing unit DS, a DFT process is executed on the
reception signal or for one symbol interval in synchronism with the time
slot signal ts, and the spectrum intensity values are calculated for the
carrier frequencies to supply the spectrum intensity output (DFT output)
dso. In the channel detection unit CD, a threshold judgment is made, per
symbol, on the spectrum intensity values of the carrier frequencies and
there are supplied the reception code data cdo corresponding to carrier
frequencies exceeding the threshold value. Here, the mode control signal
mo to be supplied from the channel detection unit CD is determined
according to the number of received channels or the number of carrier
frequencies exceeding the threshold value. Further, supplied from the
channel detection unit CD is the time slot signal ts as controlled in
phase based on the spectrum intensity values of the carrier frequencies.
In the decoding unit DE, the reception code data cdo are decoded according
to the FH or MFSK mode dependent on the state of the mode control signal
mo. After error correction, the data decoded according to the FH or MFSK
mode, are supplied as regenerated as the binary reception information data
dr.
FIG. 2 shows in detail an example of the arrangement of the transmitter T
in FIG. 1. In FIG. 2, the transmitter T comprises the coding unit CO, the
channel generation unit CG, the waveform generation unit WG, an error
correction coding unit EC, a hopping pattern generation unit HT, a mode
control unit CT, a serial-to-parallel conversion unit SP, a frequency
table CHT, a digital synthesizer ST, a mixer MXT, a reference oscillator
LT and a band-pass filter BPT. Shown in FIG. 2 are the transmission
information data dt, error correction code data eco, a hopping pattern
hto, the mode control signal mo, the transmission code data coo, S-bit
(S=log.sub.2 M) transmission code data spo, the carrier frequencies to be
used cgo, the time slot signal ts, a digital synthesizer ST output sto, a
local oscillating signal lto, a mixer MXT output mxto and the transmission
signal ot.
In FIG. 2, the coding unit CO comprises the error correction coding unit
EC, the hopping pattern generation unit HT and the mode control unit CT.
In the error correction coding unit EC, the entered transmission
information data dt are coded using a convolutional code, a block code or
the like. Generated in the hopping pattern generation unit HT is the
hopping pattern hto for a spectrum spread according to the FH mode. In the
mode control unit CT, the error correction code data eco as multiplied by
the hopping pattern hto are generated and supplied as the transmission
code data coo for the FH mode when the mode control signal mo is in the
HIGH level, and the error correction code data eco as they are, are
generated and supplied as the transmission code data coo for the MFSK mode
when the mode control signal mo is in the LOW level.
The channel generation unit CG comprises the serial-to-parallel conversion
unit SP and the frequency table CHT. In the serial-to-parallel conversion
unit SP, serially entered transmission code data coo are divided into
S-bit data and supplied in parallel as S-bit transmission code data spo
presenting information for one symbol. The frequency table CHT is formed
of a memory such as a ROM or the like containing corresponding information
of carrier frequencies for transmission code data spo, and carrier
frequencies to be used cgo are read out, from the frequency table CHT, per
entered S-bit transmission code data spo.
The waveform generation unit WG comprises the digital synthesizer ST, the
reference oscillator LT, the mixer MXT and the band-pass filter BPT. In
the digital synthesizer ST, frequency waveforms in the equivalent low band
system are digitally generated for the entered carrier frequencies to be
used cgo and supplied as hopped per symbol in synchronism with the time
slot signal ts. The reference oscillator LT generates the local
oscillating signal lto. The digital synthesizer output sto is up-converted
in frequency by the local oscillating signal lto in the mixer MXT and, in
the band-pass filter BPT, the desired band is taken out from the mixer
output mxto and supplied as the transmission signal ot to the transmission
line.
FIG. 3 shows in detail an example of the arrangement of the receiver R in
FIG. 1. In FIG. 3; the receiver R comprises the signal processing unit DS,
the channel detection unit CD, the decoding unit DE, a band-pass filter
BPR, a reference oscillator LR, a mixer MXR, a low-pass filter LPR, a
discrete Fourier transform processing unit DFT, a threshold judgment unit
3TH, a mode control signal generation unit MOG, a time slot signal
generation unit TSG, a frequency table CHR, a register RG, a mode control
unit CR, a hopping pattern generation unit HR and an error correction
decoding unit ED. Shown in FIG. 3 are the reception signal or, a band-pass
filter BPR output bpro, a local oscillating signal lro, an mixer MXR
output mxro, a low-pass filter LPR output lpro, a DFT output dso, the time
slot signal ts, carrier frequencies uno detected as channels, spectrum
intensity values uso of carrier frequencies detected as channels, the mode
control signal mo, the reception code data cdo, a register RG output rgo,
a hopping pattern hro, a hopping pattern control signal hco, a mode
control unit CR output cro and the reception information data dr.
In FIG. 3, the signal processing unit DS comprises the band-pass filter
BPR, the reference oscillator LR, the mixer MXR, the low-pass filter LPR
and the discrete Fourier transform processing unit DFT. The band-pass
filter BPR receives the reception signal or through the transmission line.
When the reception signal or is down-converted in frequency, the band-pass
filter BPR prevents the image frequency components of signals outside of
the desired band from overlapping one another. The reference oscillator LR
generates the local oscillating signal lro for down-conversion in
frequency. In the mixer MXR, the band-pass filter output bpro is
down-converted in frequency using the local oscillating signal lro. In the
low-pass filter LPR, the unnecessary signal components outside of the DFT
processing band are removed from the mixer output mxro. In the discrete
Fourier transform processing unit DFT, a DFT process is executed, in
synchronism with the time slot signal ts, on the low-pass filter output
lpro for one symbol interval and the spectrum intensity for each carrier
frequency is calculated. Here, it is required that the number of sample
points per symbol cycle in the DFT process is set to 2.times.M points or
more based on a sampling theorem and that the time slot signal ts is
accurately in synchronism with the symbol cycle.
The channel detection unit CD comprises the threshold judgment unit 3TH,
the mode control signal generation unit MOG, the time slot signal
generation unit TSG and the frequency table CHR. In the threshold judgment
unit 3TH, a threshold judgment is made, for the DFT output dso having
2.times.M points or more, on the spectrum intensity values of frequency
points corresponding to the carrier frequencies. Carrier frequencies
having spectrum intensity values exceeding the threshold value are
detected as corresponding to channels, and such spectrum intensity values
uso and such carrier frequencies uno are supplied. In the mode control
signal generation unit MOG, the channel number N (N=integer) is counted
based on the carrier frequencies detected as channels, and the mode
control signal mo is supplied in the HIGH level when N is not less than 2,
and in the LOW level when N is equal to 1. In the time slot signal
generation unit TSG, spectrum intensity values uso at consecutive symbols
are compared in level, and based on the comparison result, the time slot
signal ts is generated as controlled in phase. In the frequency table CHR,
reception code data cdo each for each carrier frequency uno are
successively read in S-bit unit from the memory table containing
corresponding information identical with that contained in the frequency
table CHT in the channel generation unit CG of the transmitter T.
The decoding unit DE comprises the register RG, the hopping pattern
generation unit HR, the mode control unit CR and the error correction
decoding unit ED. In the register RG, all reception code data cdo read out
from the frequency table CHR are updated and stored per symbol cycle, and
the register output rgo is supplied to the mode control unit CR. In the
mode control unit CR, decoding is executed with the FH mode selected when
the mode control signal mo is in the HIGH level, and with the MFSK mode
selected when the mode control signal mo is in the LOW level. In the mode
control unit CR, when the FH mode is selected, a hopping pattern control
signal hco of a channel to be received is generated such that a hopping
pattern to be followed is specified to the hopping pattern generation unit
HR. The register output rgo is inversely spread in spectrum by the hopping
pattern hro such that one of a plurality of reception code data cdo is
specified and supplied to the error correction decoding unit ED. In the
mode control unit CR, when the MFSK mode is selected, the register output
rgo is not inversely spread in spectrum but is supplied, as it is, to the
error correction decoding unit ED. In the error correction decoding unit
ED, the mode control unit CR output cro is subjected to error correction
using a convolutional code, a block code or the like and then, the
reception information data dr are regenerated and supplied.
The following description will discuss in detail the circuit operation of
the digital communication apparatus with M equal to 16 in the arrangement
in FIG. 1. It is noted that carrier frequencies are expressed in terms of
F (x) {x=1, 2, 3, . . . , 16}.
First, the operation of the transmitter T in FIG. 2 will be discussed.
FIG. 4 shows in detail an example of the arrangement of the coding unit CO
in the transmitter T in FIG. 2. In the coding unit CO in FIG. 4, the mode
control unit CT comprises a transmission rate conversion unit 4RT, an
exclusive-OR unit 4EX and a data selector 4SL. Shown in FIG. 4 are first
error correction code data eco1 and second error correction code data
eco2. The hopping pattern generation unit HT comprises a
parallel-to-serial conversion unit 4PS and a M-sequence generation unit
4MG.
The transmission information data dt are converted into the error
correction code data eco in the error correction coding unit EC and then
converted into data of the bit rate corresponding to the FH or MFSK mode
in the transmission rate conversion unit 4RT. More specifically, the first
error correction code data eco1 are supplied when the FH mode is selected,
and the second error correction code data eco2 are supplied when the MFSK
mode is selected. Since S is equal to log.sub.2 M which is equal to 4, the
second error correction code data eco2 have a bit speed four times of that
of the first error correction code data eco1. When the FH mode is
selected, the hopping pattern hto is generated in the hopping pattern
generation unit HT. In the M-sequence generation unit 4MG, therefore, a
pseudorandom code having 15 sequences is generated for every four bits and
supplied as the hopping pattern hto after converted into serial data in
the parallel-to-serial conversion unit 4PS. In the exclusive-OR unit 4EX,
the hopping pattern hto is used for a spectrum spread of the first error
correction code data eco1. Entered into the data selector 4SL are the
first error correction code data eco1, after spectrum-spread, for the FH
mode, or the second error correction code data eco2 for the MFSK mode.
When the mode control signal mo is in the HIGH level, the first error
correction code data eco1, after spectrum-spread, for the FH mode are
supplied as the transmission code data coo. When the mode control signal
mo is in the LOW level, the second error correction code data eco2 for the
MFSK mode are supplied as the transmission code data coo. The transmission
code data coo are converted into the 4-bit transmission code data spo in
the serial-to-parallel conversion unit SP in the channel generation unit
CG, and the carrier frequencies to be used cgo are read out from the
frequency table CHT containing the corresponding information shown in FIG.
5. FIG. 6 shows the contents of the frequency table CHR of the channel
detection unit CD in the receiver R corresponding to FIG. 5.
FIG. 7 shows the operational timing at the time when the FH mode is
selected in the circuit in FIG. 4, while FIG. 8 shows the operational
timing at the time when the MFSK mode is selected in the circuit in FIG.
4. In FIG. 7, carrier frequencies to be used cgo are read out in the order
of F(1).fwdarw.F(9).fwdarw.F(13) for every symbol time T. The exclusive-OR
unit 4EX in the mode control unit CR is arranged to execute an
exclusive-NOR operation between the first error correction code data eco1
and the hopping pattern hto. In FIG. 8, carrier frequencies to be used cgo
are read out in the order of F(1).fwdarw.F(2).fwdarw.F(3) for every symbol
time T.
FIG. 9 shows a frequency waveform (digital synthesizer output sto) in the
equivalent low band system for every symbol time T (t1, t2, t3) when the
MFSK mode in FIG. 8 is selected. The digital synthesizer output sto is
supplied with the hopping points thereof synchronized in phase with the
rising edges of the time slot signal ts.
FIG. 10 shows the transition of the carrier frequencies to be used in the
equivalent low band system when the MFSK mode in FIG. 8 is selected. In
FIG. 10, 16 carrier frequencies are disposed at 1/T intervals, and carrier
frequencies to be used are changed from F(1).fwdarw.F(2).fwdarw.F(3) with
the passage of time (t1, t2, t3). The digital synthesizer output sto is
up-converted in frequency to the desired band by the mixer MXT, and then
supplied to the transmission line as the transmission signal ot for
another digital communication apparatus.
The following description will discuss the operation of the receiver R in
FIG. 3.
FIGS. 11 and 12 show input timings of a DFT process in the signal
processing unit DS at the time when the receiver R in another digital
communication apparatus has received the MFSK-mode transmission signal ot
generated according to FIG. 8. FIG. 11 shows the input timings when the
DFT process intervals are not in synchronism in phase with the time slot
signal ts, while FIG. 12 shows the input timings when the DFT process
intervals are in synchronism in phase with the time slot signal ts.
In FIG. 11, a portion of the signal components of carrier frequency F(2) in
addition to a portion of the signal components of carrier frequency F(1)
is DFT-processed at a DFT process interval 1. This means that carrier
frequencies for two channels are detected for each DFT process interval.
At this time, when the level of the carrier frequency F(2) is not less
than the threshold level of the threshold judgment unit 3TH, the carrier
frequency is lowered in detection precision. This will be an obstacle to
radio communication or the like in which, for example, a distance problem
or the like is encountered. Accordingly, the time slot signal generation
unit TSG controls the phase of the time slot signal ts, thus providing
phase synchronization in FIG. 12. In the case of FIG. 10, each of the DFT
process intervals 1, 2, 3 is accurately in synchronism in phase with the
time slot signal ts. Accordingly, only one carrier frequency uno is
detected for each cycle of the time slot signal ts, thus enabling the
reception information data dr for the MFSK mode to be accurately decoded.
FIG. 13 shows channel detection in the threshold judgment unit 3TH when the
MFSK mode is selected, while FIG. 14 shows channel detection in the
threshold judgment unit 3TH when the FH mode is selected. It is now
supposed that the DFT process intervals are in synchronism in phase with
the time slot signal ts in each of FIGS. 13 and 14.
In FIG. 13, the carrier frequency F(1) which is not less than the threshold
level, is detected as corresponding to a channel, and its spectrum
intensity uso and carrier frequency uno are supplied. Carrier frequency
F(10) has more or less spectrum intensity, but its spectrum intensity is
not greater than the threshold level. Accordingly, the carrier frequency
F(10) is regarded as noise and therefore cannot be detected. In the mode
control signal generation unit MOG, the channel number N is counted as 1,
and the mode control signal mo is supplied in the LOW level.
In FIG. 14, carrier frequencies F(1), F(10), F(11) which are not less than
the threshold level, are detected as; corresponding to channels, and their
spectrum intensities uso and carrier frequencies uno are supplied. In the
mode control signal generation unit MOG, the channel number N is counted
as 3, and the mode control signal mo is supplied in the HIGH level.
FIG. 15 shows in detail an example of the arrangement of the time slot
signal generation unit TSG in the channel detection unit CD in FIG. 3. The
time slot signal generation unit TSG in FIG. 15 comprises a maximum value
detection unit MAX, a register TRG, a comparison unit COMP and a digital
variable frequency divider DVCO. Shown in FIG. 15 are maximum spectrum
intensity m.times.n, a register output m.times.p and a phase control
signal dcnt.
In FIG. 15, the maximum value detection unit MAX is arranged to detect the
maximum spectrum intensity value out of the spectrum intensity values of
the carrier frequencies detected by the threshold judgment unit 3TH, and
to supply the maximum spectrum intensity m.times.n. In the register TRG,
the maximum spectrum intensity detected earlier by one symbol time T is
stored and supplied as the register output m.times.p. In the comparison
unit COMP, the maximum spectrum intensity m.times.n and the register
output m.times.p are compared in level with each other to generate the
phase control signal dcnt. When the maximum spectrum intensity m.times.n
is greater in level than the register output m.times.p, the phase control
signal dcnt controls the phase of the digital variable frequency divider
DVCO in the same direction as that in phase control done earlier by one
symbol time T. When the maximum spectrum intensity m.times.n is smaller in
level than the register output m.times.p, the phase control signal dcnt
controls the phase of the digital variable frequency divider DVCO in the
direction opposite to the direction in phase control done earlier by one
symbol time T. This achieves the phase synchronization of the time slot
signal ts.
Using the arrangement in FIG. 1 discussed in the foregoing, there can be
achieved a highly reliable digital communication apparatus increased in
operational speed without substantial change in the hardware arrangement
of an FH-mode digital communication apparatus of prior art. In FIG. 1, the
description has been made with carrier wave transmission taken as an
example, but base band transmission may also be applied. Further, in the
spectrum analysis in the arrangement in FIG. 1, an envelop analysis using
a matched filter for each carrier frequency may also be conducted instead
of a DFT process. Such an arrangement increases the hardware size but
eliminates the need for synchronization of symbol cycle using the time
slot signal ts.
FIG. 16 shows a modification of the signal processing unit DS in FIG. 3.
The signal processing unit DS in FIG. 16 comprises a band-pass filter BPR,
a carrier sense unit EVD, a reference oscillator LR, a mixer MXR, a
low-pass filter LPR and a discrete Fourier transform processing unit DFT.
Shown in FIG. 16 are a reception signal or, a band-pass filter BPR output
bpro, a carrier sense signal evdo, a local oscillating signal lro, a mixer
MXR output mxro, a low-pass filter LPR output lpro, a DFT output dso and a
time slot signal ts.
In the carrier sense unit EVD in FIG. 16, the band-pass filter output bpro
is subjected to envelop detection to generate the carrier sense signal
evdo. In the discrete Fourier transform processing unit DFT, the low-pass
filter output lpro is subjected to the DFT process only when the carrier
sense signal evdo is asserted. This enables the discrete Fourier transform
processing unit DFT to be intermittently operated, thus lowering the power
consumption for the DFT process. Other circuit operations in the signal
processing unit DS are similar to those discussed in connection with FIG.
3.
FIG. 17 shows a modification of the coding unit CO in FIG. 2. The coding
unit CO in FIG. 17 comprises an error correction coding unit EC, a hopping
pattern generation unit HT, a mode control unit CT, a preamble generation
unit PR and a data selector 15SL. Shown in FIG. 17 are transmission
information data dt, error correction code data eco, a hopping pattern
hto, a mode control signal mo, transmission code data coo, a preamble
control signal pon and chirp code data pre.
In FIG. 17, the preamble generation unit PR generates the preamble control
signal pon for setting, as a preamble sequence, a predetermined period of
time from the point of time when the transmission information data dt have
been entered. While the preamble control signal pon is asserted, the
operation of the error correction coding unit EC is disabled and the chirp
code data pre are selected and supplied from the data selector 15SL.
Assuming that as to the chirp code data pre generated from the preamble
generation unit PR, the corresponding information is based on FIG. 5,
up-chirp code data shown in FIG. 18 and down-chirp code data shown in FIG.
19 are successively read out, per symbol time T, for up-chirp and
down-chirp, respectively. While the preamble control signal pon is being
negated, the operation of the error correction coding unit EC is enabled
and the output of the mode control unit CT is selectively supplied from
the data selector 15SL. Other circuit operations of the coding unit CO are
similar to those discussed in connection with FIG. 2.
FIG. 20 shows a modification of the receiver R in FIG. 3, corresponding to
FIG. 17. The receiver R in FIG. 20 comprises a signal processing unit DS,
a channel detection unit CD, a decoding unit DE, a register RG, a preamble
detection unit PRD, a mode control unit CR, a hopping pattern generation
unit HR and an error correction decoding unit ED. Shown in FIG. 20 are a
reception signal or, a DFT output dso, a time slot signal ts, a mode
control signal mo, reception code data cdo, an identification signal pdec,
a DFT control signal fps, an enable signal cre, a register RG output rgo,
a hopping pattern hro, a hopping pattern control signal hco, a mode
control unit CR output cro and reception information data dr.
In the preamble detection unit PRD in FIG. 20, a preamble using an up-chirp
is identified by following carrier frequency while successively changing,
per symbol time T, the carrier frequencies from the least significant
carrier frequency F(1) to the most significant carrier frequency F(16)
from the point of time where the register output rgo coincides with the
reception code data cdo corresponding to the least significant carrier
frequency F(1). On the other hand, a preamble using a down-chirp is
identified by following carrier frequency while successively changing, per
symbol time T, the carrier frequencies from the most significant carrier
frequency F(16) to the least significant carrier frequency F(1) from the
point of time where the register output rgo coincides with the reception
code data cdo corresponding to the most significant carrier frequency
F(16). In the preamble detection unit PRD, when the preamble
identification is determined, the enable signal cre is supplied in
synchronism with completion of the preamble to cause the mode control unit
CR to start controlling the phase of the hopping pattern hro to be used.
In the preamble detection unit PRD, when the preamble identification is
established, the identification signal pdec is supplied to the threshold
judgment unit 3TH (See FIG. 3) in the channel detection unit CD. In the
threshold judgment unit 3TH, when the identification signal pdec is
asserted, the DFT control signal fps controls the operation of the
discrete Fourier transform processing unit DFT (See FIG. 3) in the signal
processing unit DS.
The following description will discuss in detail the operational control
(frequency error correction control) for the discrete Fourier transform
processing unit DFT in the arrangements in FIGS. 17 and 20.
It is now supposed that, in the digital synthesizer ST of the transmitter T
at the preamble sequence, there are successively generated the signal
waveforms of carrier frequencies corresponding to the up-chirp signals
(F(1).fwdarw.(F16)) or down-chirp signals (F(16).fwdarw.F(1)) in FIG. 21.
In FIG. 21, f(p) {P=0, 1, 2, . . . , 127} refers to the frequency point
scaled at equal intervals on the frequency coordinates on the basis of the
transmitter T. According to the following formula (1), F(1) generated in
the digital synthesizer ST corresponds to the frequency point f(8), F(2)
corresponds to the frequency point f(10) and F(16) corresponds to the
frequency point f(38) :
F(x).fwdarw.f(2x+6) {x=1, 2, 3, . . . , 16} (1)
In another digital communication apparatus, the discrete Fourier transform
processing unit DFT of the signal processing unit DS in the receiver R
executes, on the low-pass filter output lpro after down-converted in
frequency, a DFT process with frequency resolution of carrier frequency
distance 1/(2T). It is now supposed that there is a frequency error
.DELTA.fa between the frequency of the local oscillating signal lto of the
transmitter T and the local oscillating signal lro of the receiver R. In
such a case, a DFT process is executed, on the reference frequency
coordinates, on the least significant carrier frequency F(1) as shown in
FIG. 22 and on the most significant carrier frequency F(16) as shown in
FIG. 23. At this time, when the spectrum intensity for the least
significant carrier frequency F(1) is not related to the frequency point
f(8) but is related to the frequency point f(9), and when the spectrum
intensity for the most significant carrier frequency F(16) is not related
to the frequency point f(38) but is related to the frequency point f(39),
the frequency error becomes .DELTA.fb smaller than .DELTA.fa and is nearer
to the actual value. It is now supposed that the variable frequency range
width at the preamble sequence is previously known and that the frequency
error .DELTA.fa is smaller than the channel interval 1/T. In this case, by
detecting the frequency point of the maximum spectrum intensity for the
least significant carrier frequency F(1) or the most significant carrier
frequency F(16), it becomes possible to correct a frequency error between
a plurality of digital communication apparatus. For example, in the case
of FIGS. 22 and 23, carrier frequencies in the threshold judgment unit 3TH
(See FIG. 3) of the channel detection unit CD may be changed for the
frequency point expressed by the following formula (2):
F(x).fwdarw.f(2x+7) {x=1, 2, 3, . . . , 16} (2)
Thus, using the arrangements in FIGS. 17 and 20, a preamble for up-chirt or
down-chirp can be generated and detected with the use of a simple circuit
arrangement. Further, the use of such a preamble facilitates carrier sense
in the multipath fading environment. In the arrangement in FIG. 20,
frequency error correction is made only when the preamble is detected and,
after the completion of the preamble, a partial DFT process with frequency
resolution of 1/T is executed on channels after frequency error
correction. This not only improves the reception sensibility but also
reduces the operations to be executed in amount.
The following description will discuss modifications of the contents stored
in the frequency tables in FIGS. 5 and 6. FIG. 24 shows information stored
in the frequency table CHT of the channel generation unit CG (See FIG. 2),
while FIG. 25 shows information stored in the frequency table CHR of the
channel detection unit CD (See FIG. 3). In this modification, the hardware
structure is the same as that discussed in connection with FIGS. 1 to 3.
In this modification, 4-bit transmission code data spo and carrier
frequencies to be used cgo in FIG. 24 correspond to each other using a
Gray code as the progressive code, and carrier frequencies uno and 4-bit
reception code data cdo in FIG. 25 correspond to each other using a Gray
code as the progressive code. Since the progressive code is used for
arrangement of carrier frequencies, adjacent carrier frequencies undergo a
change in bit information only by one bit. This lowers the influence of
error detection due to frequency error among a plurality of digital
communication apparatus.
When using the progressive code and corresponding information in FIGS. 24
and 25, the chirp code data in FIG. 26 may be used for generating an
up-chirp signal, and the chirp code data in FIG. 27 may be used for
generating a down-chirp signal.
FIG. 28 shows an example of the arrangement of the digital communication
apparatus according to the present invention. The apparatus in FIG. 28
comprises a transmitter T, a receiver R, a frequency selection unit CE, a
waveform generation unit WG, a cosine-wave and sine-wave generation unit
WCS, a cosine-wave generation unit WC, a sine-wave generation unit WS, a
mixer WCM, a mixer WSM, a 90.degree. phase shifter WP, an adder WA, a 1/2
frequency divider DV, an oscillator SG, a down-converter unit FD, a DFT
operation unit DP, a threshold judgment unit 28CT, a synchronizing signal
generation unit SC, a latch unit 28LT and a decoder 28CD. Shown in FIG. 28
are transmission data dt, carrier frequencies to be used uc, a cosine-wave
generation unit WC output wco, a sine-wave generation unit WS output wso,
a transmission signal wt, a system clock sysc with frequency 2/.DELTA.t, a
sampling clock smpc with frequency 1/.DELTA.t, a reference oscillating
signal sgo with frequency fc, a reception signal wr, a down-converter unit
FD output wrd, a DFT operation unit DP spectrum value output I(k) dpo, a
threshold judgment unit 28CT spectrum value output I(k) and candidate
carrier frequency cto, a synchronizing trigger signal st, reception
carrier frequencies 28lto and reception data dr. Here, .DELTA.t refers to
the time period of the sampling clock smpc. The transmitter T and the
receiver R share the same 1/2 frequency divider DV and the same oscillator
SG.
In FIG. 28, binary transmission data dt which are coded according to the FH
or MFSK mode outside of the apparatus, are serially entered into the
transmitter T. In the frequency selection unit CE, serial transmission
data dt are divided into blocks each having log.sub.2 M bits for every two
time slots, and for each block, corresponding carrier frequencies to be
used uc out of the carrier frequencies F(k) (k=1, 2, . . . , M) are read
out from the internal table; M is an integer not less than 2 and
represents the number of carrier frequencies to be used or channels to be
used, i.e., the series length of transmission data. In the waveform
generation unit WG, the frequency waveforms for the carrier frequencies to
be used uc are digitally generated and are supplied, to the transmission
line, as the transmission signal wt in synchronism with the time slot by
the synchronizing trigger signal st generated in the receiver R.
More specifically, according to the value of k in F(k) of the carrier
frequencies to be used uc, the cosine-wave generation unit WC and the
sine-wave generation unit WS in the waveform generation unit WG
respectively digitally generate, in synchronism with the synchronizing
trigger signal st generated in the receiver R, cosine waves and sine waves
to be used for a frequency orthogonal transformation. It is now supposed
that there is used a frequency orthogonal transformation represented by
the following equation (3):
##EQU1##
Then, cosine waves represented by
(2.pi..times..DELTA.f.times.(2k-1).times.t) and sine waves represented by
(-1).sup.k-1 .times.sin (2.pi..times..DELTA.f.times.(2k-1).times.t) are
respectively generated by the cosine-wave generation unit WC and the
sine-wave generation unit WS. In the equations above-mentioned, .DELTA.f
refers to a frequency step width and t refers to time. In the two mixers
WCM, WSM and in the adder WA, there is conducted, using cosine waves and
sine waves, a frequency orthogonal transformation on the reference
oscillating signal sgo from the oscillator SG, i.e., cos
(2.pi..times.fc.times.t), and on the signal from the 90.degree. phase
shifter WP, i.e., sin (2.pi..times.fc.times.t). Thus, in synchronism with
the time slot, the frequency waveforms for the carrier frequencies to be
used uc are generated and supplied as the transmission signal wt to the
transmission line. Instead of the equation (3), the following equation (4)
may also be used:
##EQU2##
The reception signal wr is entered into the receiver R through the
transmission line. In the down-converter unit FD, signal components in the
variable frequency band to be received have been taken from the reception
signal wr, and the signal components thus taken are down-converted in
frequency, using the reference oscillating signal sgo from the oscillator
SG, toward a low frequency band in which a digital signal process can be
executed. In the DFT operation unit DP, an N-point DFT process for a
period of the latest one time slot (T=N.times..DELTA.t) is successively
executed on the down-converter output wrd for each sampling clock cycle
.DELTA.t. Here, N is an integer not less than M. At this time, the DFT
process is executed only on M frequency points where each carrier
frequency is mapped. Accordingly, the spectrum value I(k) is calculated
for each carrier frequency F(k) (k=1, 2, . . . , M) per sampling clock. In
the threshold judgment unit 28CT, a threshold judgment is made on the
spectrum value I(k) for each channel using a level determined by a
spurious response. Then, each carrier frequency exceeding a threshold
value TH is supplied, as a candidate carrier frequency Fd(l) (l=0, . . . ,
s), together with the spectrum value I(k) of each carrier frequency. In
the synchronizing signal generation unit SC, the time slot is
synchronously extracted based on the threshold judgment unit output cto,
i.e., the spectrum values I(k) and the candidate carrier frequencies
Fd(l), thus generating the synchronizing trigger signal st. In the latch
unit 28LT, the candidate carrier frequencies Fd(l) at the time when the
synchronizing trigger signal st has been asserted, are latched and
supplied as the reception carrier frequencies 28lto. In the decoder unit
28CD, data of a block having log.sub.2 M bits corresponding to reception
carrier frequencies 28lto are read out from the internal table and
supplied as the reception data dr. The reception data dr are decoded,
outside of the apparatus, according to the FH or MFSK mode.
FIG. 29 shows carrier frequencies so arranged as to correspond to 4-bit
data with M being equal to 16, using a Gray code as the progressive code
in the frequency selection unit CE and the decoder unit 28CD in FIG. 28.
Based on FIG. 29, it is possible to reduce the error to one bit or less
when 4-bit reception data dr based on the reception carrier frequencies
28lto are decoded, as far as the frequency conversion error is within one
carrier frequency interval (2.times..DELTA.f).
The following description will discuss in detail an example in which M is
equal to 16, N is equal to 64, .DELTA.t is equal to 1/16 (.mu.s) and R is
equal to 1 in the arrangement in FIG. 28. Here, a period of one time slot
T is equal to N.times..DELTA.t which is equal to 4 .mu.s, and the
frequency step width .DELTA.f is as follows:
.DELTA.f=1/T.times.R=1/(N.times..DELTA.t)=250 kHz
FIG. 30 shows in detail an example of the arrangement of the cosine-wave
and sine-wave generation unit WCS in FIG. 28. In FIG. 30, the cosine-wave
and sine-wave generation unit WCS comprises a 7-bit accumulator 2AC, an
inverter 2IV, a data selector 2MP, a cosine-wave memory 2CM, a sine-wave
memory 2SM, D/A converters 2CDA, 2SDA, and low-pass filters 2CLF, 2SLF.
Shown in FIG. 30 are the synchronizing trigger signal st from the receiver
R, the system clock sysc; an accumulator output 2aco, an inverter output
2ivo, carrier frequencies to be used uc of F(k) (k=1, 2, . . . , 16) given
from the frequency selection unit CE, cosine-wave data 2cd, sine-wave data
2sd, the cosine-wave generation unit output wco and the sine-wave
generation unit output wso.
In FIG. 30, based on the value of k in F(k) (k=1, 2, . . . , 16) of the
carrier frequencies to be used uc determined for each time slot, the
accumulator 2AC generates, for each system clock sysc (of which frequency
is equal to 2/.DELTA.t or 32 MHz), the accumulator output 2aco while the
values of 2k-1 are successively accumulated in a binary operation. Based
on the assumption that waveform data for 2.times.N=128 (=2.sup.7) points
are read out per time slot, the accumulator output 2aco has a 7-bit width
(0000000.about.1111111), and is circulatingly operated for every overflow
and reset each time the synchronizing trigger signal st is asserted. In
the cosine-wave memory 2CM and the sine-wave memory 2SM, cosine-wave data
in which one cycle is being sampled to 128 points and sine-wave data in
which one cycle is being sampled to 128 points, are respectively stored,
as quantized, up to the address of 1111111 with the address 0000000
serving as the phase 0.degree.. In the cosine-wave memory 2CM, the
cosine-wave data 2cd are read out with the accumulator output 2aco used as
an address. In the sine-wave memory 2SM, based on the data selector 2MP,
the sine-wave data 2sd are read out according to the equation (3)
dependent on k which is an odd or even number. That is, when k is an odd
number, the sine-wave data 2sd are read out with the accumulator output
2aco used as an address, and when k is an even number, the sine-wave data
2sd are read out using, as an address, the inverter output 2ivo in which
the polarity of the accumulator output 2aco is being inverted by the
inverter 2IV. When conducting the frequency orthogonal transformation
represented by the equation (4), there are executed, according to k which
is an even or odd number, the operations inverse to those above-mentioned.
By the two D/A converters 2CDA, 2SDA, the read cosine-wave data 2cd and
sine-wave data 2sd are converted into analog signals which are in
synchronism with the system clock sysc. These analog signals are smoothed
by the two low-pass filters 2CLF, 2SLF such that there are generated cos
(2.pi..times..DELTA.f.times.(2k-1).times.t) as the cosine-wave generation
unit output wco and (-1).sup.k-1
.times.sin(2.times..DELTA.f.times.(2k-1).times.t) as the sine-wave
generation unit output wso.
FIG. 31 shows the arrangement of M (=16) frequencies after frequency
orthogonal transformation represented by the equation (3). based on the
values of k in the carrier frequencies F(k), the frequencies are
alternately arranged, on the basis of the frequency fc, toward the high
band side by .DELTA.f.times.(2k-1) when k is an odd number, and toward the
low band side by .DELTA.f.times.(2k-1) when k is an even number.
FIG. 32 shows the arrangement of M (=16) frequencies after frequency
orthogonal transformation represented by the equation (4). Based on the
values of k in the carrier frequencies F(k), the frequencies are
alternately arranged, on the basis of the frequency fc, toward the low
band side by .DELTA.f.times.(2k-1) when k is an odd number, and toward the
high band side by .DELTA.f.times.(2k-1) when k is an even number.
As shown in FIGS. 31 and 32, the carrier frequencies are alternately
arranged with respect to the frequency fc serving as the basis.
Accordingly, even though a spurious response is generated in side bands
due to normalization level error between the sine-wave generation unit
output wso and the cosine-wave generation unit output wco at the frequency
orthogonal transformation (orthogonal modulation), the influence exerted
upon other carrier frequencies can be reduced.
FIG. 33 shows the arrangement of carrier frequencies which will be mapped
in frequency when the carrier frequencies F(k) (k=1, 2, . . . , M) after
frequency orthogonal transformation shown in FIG. 31, are down-converted
in frequency, using the reference oscillating signal sgo having the
frequency fc (fc to DC), by the down-converter unit FD of the receiver R.
After the carrier frequencies have been down-converted in frequency, the
frequency interval is changed from 4.times..DELTA.f to 2.times..DELTA.f
and therefore the occupied frequency bandwidth becomes a half of the
variable frequency range. In the DFT operation unit DP, therefore, a DFT
process can be executed using frequency of 16.times.4.times..DELTA.f equal
to the variable frequency range, i.e., 16 MHz, as sampling frequency fs.
Instead of the frequency orthogonal transformation represented by the
equation (3), there may be used a frequency orthogonal transformation
represented by the following equation (5):
##EQU3##
Instead of the frequency orthogonal transformation represented by the
equation (4), there may be used a frequency orthogonal transformation
represented by the following equation (6):
##EQU4##
Except for a binary number operation of the accumulator 2AC shown in FIG.
30, the hardware arrangements are the same. That is, according to the
values of k in the carrier frequencies F(k), k is successively
accumulated.
FIG. 34 shows the arrangement of M (=16) carrier frequencies after
frequency orthogonal transformation represented by the equation (5). Based
on the values of k in the carrier frequencies F(k), the frequencies are
alternately arranged, on the basis of the frequency fc, toward the high
band side by .DELTA.f.times.k when k is an odd number, and toward the low
band side by .DELTA.f.times.k when k is an even number.
FIG. 35 shows the arrangement of M (=16) carrier frequencies after
frequency orthogonal transformation represented by the equation (6). Based
on the values of k in the carrier frequencies F(k), the frequencies are
alternately arranged, on the basis of the frequency fc, toward the low
band side by .DELTA.f.times.k when k is an odd number, and toward the high
band side by .DELTA.f.times.k when k is an even number.
Even though the equation (5) or (6) is used, there can be produced effects
similar to those discussed in connection with FIGS. 31 and 32. Further,
the carrier frequency interval can be narrower than in FIGS. 31 and 32. It
is therefore possible to provide a larger number of carrier frequencies.
It is however noted that the influence of a spurious response among the
carrier frequencies becomes greater.
FIG. 36 shows in detail an example of the arrangement of the DFT operation
unit DP (k) for one carrier frequency F(k) in the DFT operation unit DP in
FIG. 28. Shown in FIG. 36 are a DFT operation unit DP(k) for the carrier
frequency F(k), an A/D converter 8AD, a log.sub.2 N-bit counter 8CO, a
cosine-wave memory 8CM, a sine-wave memory 8SM, a data selector 8MP, a
multiplier 8MX, an a-stage 2.times.N-bit shift register 8SFA, an
arithmetic operation unit (ALU) 8AL, a b-stage 2-bit shift register 8SFB,
an absolute value operation unit (ABS) 8AB and a flip-flop 8FF. Here,
log.sub.2 N is equal to 6, and 2.times.N.sup.- is equal to 128. Also
shown in FIG. 36 are a down-converter output wrd, a system clock sysc (of
which frequency is equal to 2/.DELTA.t or 32 MHz), a sampling clock smpc
(of which frequency is equal to 1/.DELTA.t or 16 MHz), a log.sub.2 N-bit
counter 8CO output 8coo, an A/D converter 8AD output 8ado, cosine-wave
data 8cmo, sine-wave data 8smo, a data selector 8MP output 8mpo,
a-multiplier 8MX output 8mxo, a 2.times.N shift register 8SFA output
8sfao, an arithmetic operation unit 8AL output 8alo, a 2-bit shift
register 8SFB output 8sfbo, an absolute value operation unit 8AB output
8abo and a spectrum value I(k) for the carrier frequency F(k). Further
shown in FIG. 36 are a 2-bit shift register 8SFB's first bit output 8sfb1
and a 2-bit shift register 8SFB's second bit output 8sfb2.
In the DFT operation unit DP(k) for the carrier frequency F(k), when the
down-converter output value is defined as Wd, the spectrum value I(k) is
calculated by executing the following operations of equation (7) per k:
##EQU5##
in which sqrt () means a square root function and p means the absolute
time point at this point of time.
In FIG. 36, the down-converter output wrd is converted into an a-bit
digital signal by the A/D converter 8AD per sampling clock smpc or per j
in the equations (7). Cosine-wave data and sine-wave data for N (=64)
points having a time length of one time slot corresponding to the carrier
frequency F(k), are respectively stored in the cosine-wave memory 8CM and
the sine-wave memory 8SM, and are read out, per sampling clock smpc, using
the log.sub.2 N-bit counter 8CO output 8coo as an address. The data
selector 8MP is arranged to execute a process using, in time-division
multiplexing, the hardware for partial operations (multiplication,
addition and subtraction) in the DFT process, and to alternately assign
the cosine-wave data 8cmo and the sine-wave data 8smo to the multiplier
8MX per system clock sysc. In the multiplier 8MX, multiplication of
Wd(j).times.cos (2.pi..times..DELTA.f.times.(2k-1).times..DELTA.t.times.j)
in the equation (7) and multiplication of Wd(j).times.sin
(2.pi..times..DELTA.f.times.(2k-1).times..DELTA.t.times.j) in the equation
(7), are alternately executed for the A/D converter 8AD a-bit output 8ado
and the cosine-wave data 8cmo, and for the A/D converter 8AD a-bit output
8ado and the sine-wave data 8smo, thus supplying the upper a-bit
multiplication result 8mxo. Each of the a-stage 2.times.N-bit shift
register 8SFA and the b-stage 2-bit shift register 8SFB, is operated at
the system clock sysc. This provides a delay of the multiplication result
8mxo by a period of one time slot in the 2.times.N shift register 8SFA and
a delay of the arithmetic operation unit output 8alo by a period of one
sampling clock in the 2-bit shift register 8SFB. In the arithmetic
operation unit 8AL, the shift register output 8sfbo is added to the
multiplication result 8mxo per system clock sysc, and the shift register
output 8sfao is subtracted from the multiplication result 8mxo per system
clock sysc, thus calculating and supplying Ic(j) or Is(j) which is the
b-bit accumulation result (correlation value) for a period of the latest
one time slot. In the absolute value operation unit 8AB, there is
calculated, per system clock sysc, the square mean (sqrt (Ic(j).sup.2
+Is(j).sup.2) or (sqrt (Is(j).sup.2 +Ic(j+1).sup.2) of the 2-bit shift
register 8SFB's first bit output 8sfb1 and the 2-bit shift register 8SFB's
second bit output 8sfb2, and the absolute value operation unit output 8abo
at the same timing j is supplied as the spectrum value I(k) from the
flip-flop 8FF.
In FIG. 28, the DFT operation unit DP is formed of 16 DFT operation units
DP(k) each shown in FIG. 36, and other arrangement than the 16 DFT
operation units DP(k) is common for different k. Further, the sampling
frequency for the DFT process is advantageously reduced to 1/2. In FIG.
36, therefore, the hardware is reduced in size due to the time-division
arrangement of the multiplier 8MX and the arithmetic operation unit 8AL.
As to the absolute value operation unit 8AB, too, a time-division
arrangement can be used between two carrier frequencies, for example,
k=(1, 2), k=(3, 4), k=(5, 6), k=(7, 8), k=(9, 10), k=(11, 12), k=(13, 14),
k=(15, 16).
FIG. 37 shows in detail an example of the arrangement of the threshold
judgment unit 28CT in FIG. 28. Shown in FIG. 37 are a threshold value
control unit GT and the kth comparator 9C(k) (k=1, 2, . . . , 16) for the
carrier frequency F(k). Also shown in FIG. 37 are a spectrum value I(k)
corresponding to the carrier frequency F(k), a sampling clock smpc (of
which frequency is equal to 1/.DELTA.t or 16MHz), a threshold value TH and
the kth enable signal en(k) corresponding to the carrier frequency F(k).
In FIG. 37, each spectrum value I(k) calculated in each DFT operation unit
DP(k) in FIG. 36 is compared in each comparator 9C(k) with the threshold
value TH set by the threshold value control unit GT according to the
spurious level at the time of DFT process. Then, there is generated each
enable signal en(k) corresponding to each carrier frequency F(k). When the
spectrum value I(k) does not exceed the threshold value TH, the enable
signal en(k) is asserted in the HIGH level, and when the spectrum value
I(k) exceeds the threshold value TH, the enable signal en(k) is negated in
the LOW level. From the threshold judgment unit 28CT, 16 enable signals
en(k) and 16 spectrum values I(k) are simultaneously supplied as threshold
judgment unit output cto (K=1, 2, . . . , 16), and the spectrum value of
each carrier frequency negated in the LOW level is set as the spectrum
value Id(l) (l=0, . . . , s) of each candidate carrier frequency.
FIG. 38 shows in detail an example of the arrangement of the threshold
value control unit GT in FIG. 37. Shown in FIG. 38 are a maximum value
detection unit 10MD, a comparator 10CP, a maximum value register 10RM, a
divider 10DV, an integrator 10I and a threshold value memory table 10MT.
Also shown in FIG. 38 are a spectrum value I(k) corresponding to each
carrier frequency F(k) (k=1, 2, . . . , 16), a sampling clock smpc (of
which frequency is equal to 1/.DELTA.t or 16 MHz), a maximum value
detection unit 10MD output 10mdo, a comparator 10CP output 10cpo, a
register 10RM output 10rmo, a divider 10DV output 10dvo, an integrator 10I
output 10io and a threshold value TH.
In FIG. 38, spectrum values I(1).about.I(16) are entered into the maximum
value detection unit 10MD per sampling clock smpc, and the maximum
spectrum value Im1 is selected and supplied as the maximum value detection
unit output 10mdo. In the comparator 10CP, the previous maximum value Im
which is the maximum register 10RM output 10rmo, is compared in level with
the new maximum value Im1. Only when Im1 is greater than Im, the
comparator output 10cpo is asserted and IM1 is latched in the register
10RM. In the divider 10DV, normalization is made by an operation of
Im1/Im. The divider output 10dvo is smoothed by the integrator 10I and
supplied as the integrator output 10io which serves as an address for the
threshold value memory table 10MT. From the threshold value memory table
10MT, the threshold value TH is selected and read out.
The threshold value control unit GT in FIG. 38 is advantageous when an
automatic gain control (AGC) is made in the down-converter unit FD. When
the maximum AGC output is in a predetermined level, the spectrum value of
each candidate carrier frequency is smaller as the multiplicity of the
transmission channel number s is greater. Accordingly, when the threshold
value TH is set in association with the integrator output 10io, the
optimum threshold value TH for the transmission channel number s is
selected, thus improving the carrier frequency detection precision.
FIG. 39 shows in detail an example of the arrangement of the synchronizing
signal generation unit SC in FIG. 28. Shown in FIG. 39 are the kth data
selector 11 MP(k) (k=1, 2, . . . , 16) for the carrier frequency F(k), a
total sum operation unit 11S, an arithmetic operation unit (ALU) 11AL, an
N.times.TC-bit shift register 11SFA, an N-bit shift register 11SFB, a
minimum value detector unit 11MD, a first comparator 11CP, an A register
11RA, a B register 11RB, a C register 11RC, a log.sub.2 N-bit counter
11CO, a second comparator 11DE and a clock regeneration unit CR. Here, TC
(which is an integer not less than 1) represents time constant, and
log.sub.2 N is equal to 6. In FIG. 39, cto generally designates both
spectrum value output I(k) and enable signal en(k) (k=1, 2, . . . , 16)
which are given from the threshold judgment unit 28CT in FIG. 37. Also
shown in FIG. 39 are a sampling clock smpc (of which frequency is equal to
1/.DELTA.t or 16 MHz), a total sum operation unit 11S output 11so, an
arithmetic operation unit 11AL output 11alo, an N.times.TC-bit shift
register 11SFA output 11sfao, an N-bit shift register 11SFB output 11sfbo,
an A register 11RA output 11rao, a first comparator 11CP output 11cpo, a B
register 11RB output 11rbo, a C register 11RC output 11rco, a log.sub.2
N-bit counter 11CO count output 11cuo, a log.sub.2 N-bit counter 11Co
carry output 11car, a second comparator 11DE output deo and a
synchronizing trigger signal st.
In FIG. 39, whether or not the spectrum values I(1) to I(16) supplied from
the threshold judgment unit 28CT are to be entered into the total sum
operation unit 11S, is selected, in the 16 data selectors 11MP(1) to
11MP(16), using the enable signals en(1) to en(16) supplied from the
threshold judgment unit 28CT. That is, there is selected only the spectrum
value I(k) of each carrier frequency of which enable signal en(k) is
asserted in the HIGH level, i.e., only the spectrum value I(k) of carrier
frequency other than the candidate carrier frequency Id(l) (l=0, 1, . . .
, s). Each spectrum value I(k) thus selected is entered and added in the
total sum operation unit 11S. Then, there is operated a cost function Cf
of the following equation (8):
Cf=.SIGMA..sub.k=1.sup.16 (I(k))-.SIGMA..sub.l=0.sup.s (Id(l))(8)
Per sampling clock smpc, the total sum operation unit 11S supplies a cost
function Cf as the total sum operation unit output 11so. Both the
N.times.TC-bit shift register 11SFA and the N-bit shift register 11SFB are
operated at the sampling clock smpc. The total sum operation unit output
11so is delayed by a period of TC time slots in the N.times.TC-bit shift
register 11SFA, and the arithmetic operation unit output 11alo is delayed
by a period of one time slot in the N-bit shift register 11SFB. In the
arithmetic operation unit 11AL, the total sum operation unit output 11so
is subjected to addition with respect to the N-bit shift register output
11sfbo, and is subjected to subtraction with respect to the N.times.TC-bit
shift register output 11sfao. Per time point i (i=1, 2, . . . N) in
.DELTA.t of time slot, a cost function accumulated value C(i) for time
constant TC is calculated and supplied as the arithmetic operation unit
output 11alo. In the minimum value detection unit 11MD, the A register
output 11rao which is a temporary candidate of the minimum cost function
accumulated value C(i) in a period of one time slot, is compared with the
N-bit shift register output 11sfbo in the first comparator 11CP. Only when
the N-bit shift register output 11sfbo is smaller, the first comparator
output 11cpo is asserted in the HIGH level and the N-bit shift register
output 11sfbo is latched in the A register 11RA. In the log.sub.2 N-bit
counter 11CO, counting-up is made per sampling clock smpc to count a
period of one time slot (N=64 points). The count output 11cuo at the time
when the first comparator output 11cpo has been asserted, is latched in
the B register 11RB. Accordingly, when the carry output 11car is asserted,
the timing (time point i) at which the cost function accumulated value
C(i) is minimized within one time slot, is determined based on the B
register output 11rbo, and the B register output 11rbo at this time is
latched in the C register 11RC. Simultaneously, the A register output
11rao is preset to 1 by the carry output 11car for detection the minimum
value within a period of a subsequent time slot. In the second comparator
11DE, whether or not the C register output 11rco coincides with the count
output 11cuo, is detected within a period of a subsequent time slot. When
a coincidence is detected, the second comparator output deo is supplied as
asserted in the HIGH level. In the clock regeneration unit CR, the time
slot for the second comparator output deo is stabilized, thereby to supply
the synchronizing trigger signal st.
FIG. 40 shows in detail an example of the arrangement of the clock
regeneration unit CR in FIG. 39. Shown in FIG. 40 are a first comparator
12CP1, a log.sub.2 N-bit counter 12CO, a data selector 12MP, a second
comparator 12CP2, an up/down counter 12CUD and an inverter 12IV. Here,
log.sub.2 N is equal to 6. Also shown in FIG. 40 are the output deo of the
second comparator 11DE in the synchronizing signal generation unit SC in
FIG. 39, a first comparator 12CP1 output 12cp1o, sampling clock smpc (of
which frequency is equal to 1/.DELTA.t or 16 MHz), a log.sub.2 N-bit
counter 12CO output 12coo, a data selector 12MP output 12mpo, an up/down
counter 12CUD output 12cudo, an upper limit value 12svu for the second
comparator 12CP2, a lower limit value 12svd for the second comparator
12CP2, a second comparator 12CP2 phase output 12cdp, a second comparator
12CP2 detection output 12cdt, the most significant bit (MSB) 12msb of the
log.sub.2 N-bit counter output 12coo and the synchronizing trigger signal
st.
In the log.sub.2 N-bit counter 12CO in FIG. 40, counting-up from 000000 to
111111 is repeated per sampling clock smpc. At this time, the most
significant bit (MSB) 12msb of the counter becomes the synchronizing
trigger signal st through the inverter 12IV. In the first comparator
12CP1, the counter output 12coo at the timing where the input deo is
asserted, is compared with a 6-bit data value 000000 such that the phase
information of the synchronizing trigger signal st with respect to the
time slot, is detected. More specifically, the first comparator output
12cp1o is supplied on the assumption that the phase is led while the
counter output 12coo has a value from 000001 to 011111, and that the phase
is delayed while the counter output 12coo has a value from 100000 to
111111. In the up/down counter 12CUD, based on the first comparator output
12cp1o, counting-up is made when the phase is led, and counting-down is
made when the phase is delayed. In the second comparator 12CP2, a
threshold judgment is made, based on the upper and lower limit values
12svu, 12svd, on the up/down counter output 12cudo integrated per time
slot. When the up/down counter output 12cudo deviates from the upper or
lower limit value, there are generated a detection output 12cdt and a
phase output 12cpd for phase correction. More specifically, when the
up/down counter output 12cudo exceeds the upper limit value 12svu, the
phase output 12cdp causes the data selector 12MP to select, at the timing
of the counter output 12coo of 000000, the data value 111111, and this
data value 111111 is loaded on the log.sub.2 N-bit counter 12CO, such that
a 1-bit lead correction is made on the phase. On the other hand, when the
up/down counter output 12cudo is below the lower limit value 12svd, the
phase output 12cdp causes the data selector 12MP to select, at the timing
of the counter output 12coo of 000000, the data value 000001, and this
data value 000001 is loaded on the log.sub.2 N-bit counter 12CO, such that
a 1-bit lag correction is made on the phase. Each time phase correction is
made, the up/down counter output 12cudo is reset by the detection output
12cdt. The operations above-mentioned are successively repeated such that
the synchronizing trigger signal st is accurately in synchronism with the
time slot.
FIG. 41 shows a timing chart illustrating the input/output relation in the
clock regeneration unit CR in FIG. 40. Even though a jitter is generated
in the input deo, a stable synchronizing trigger signal st is acquired
because of the integration effect produced by the up/down counter 12CUD.
It is assumed that the synchronizing trigger signal st is asserted at its
rising edges.
FIG. 43 shows, for 8 time slots (8.times.256=2048 points), the cost
function accumulated values C(i) for time constant TC=16 time slots in the
synchronizing signal generation unit SC in FIG. 39 when M is equal to 16
and N is equal to 256 and when two channels shown in FIG. 42 are received.
FIG. 45 is a view similar to FIG. 43 at the time when two channels shown
in FIG. 44 are received. In each of FIGS. 43 and 45, no noise is being
added.
In FIG. 43, there are obtained cost function accumulated values C(i) broad
in dynamic range because the carrier frequencies in the reception channel
1 and the reception channel 2 do not overlap each other in the zone
extending over two time slots in FIG. 42 such that the randomized property
is high. In FIG. 45, however, there are only obtained function accumulated
values C(i) narrow in dynamic range because the carrier frequencies in the
reception channel 1 and the reception channel 2 overlap each other in the
zone extending over two time slots in FIG. 44 such that the randomized
property is low.
FIG. 47 shows, for 8 time slots (8.times.256=2048 points), cost function
accumulated values C(i) for time constant TC=16 time slots in the
synchronizing signal generation unit SC in FIG. 39 when M is equal to 16
and N is equal to 256 and when three channels shown in FIG. 46 are
received. FIG. 49 is a view similar to FIG. 47 at the time when three
channels in FIG. 48 are received. In each of FIGS. 47 and 49, no noise is
being added.
In FIGS. 47 and 49 in which the number of reception channels is three, too,
there are obtained the results of cost function accumulated values C(i)
similar to those obtained when the number of reception channels is two.
When the transmission data are enhanced in randomized property, there are
obtained cost function accumulated values C(i) broad in dynamic range.
This means that when frequency hopping is made using a Reed-Solomon code
or the like, cost function accumulated values C(i) broad in dynamic range
are obtained as an inevitable consequence.
FIG. 51 shows, for 8 time slots (8.times.256 2048 points), cost function
accumulated values C(i) for time constant TC=1 time slot in the
synchronizing signal generation unit SC in FIG. 39 when M is equal to 16
and N is equal to 256 and when one channel shown in FIG. 50 is received.
FIG. 52 shows, for 8 time slots (8.times.256=2048 points), cost function
accumulated values C(i) for time constant TC=16 time slots in the
synchronizing signal generation unit SC in FIG. 39 when M is equal to 16
and N is equal to 256 and when one channel shown in FIG. 50 is received.
In each of FIGS. 51 and 52, noise (S/N=6 dB) is being added.
As apparent from FIG. 52, on the assumption that, under the environment
where noise is present, frequency hopping is conducted using random
transmission data having a series length of M, when the time constant TC
is set to a value equal to the product of the number of carrier
frequencies M and a positive integer, the generated cost function
accumulated values C(i) are stabilized by sufficient averaging, thus
stabilizing the synchronizing trigger signal st.
FIG. 53 shows an example of the arrangement of the digital communication
apparatus according to the present invention, in which the oscillator SG
in FIG. 28 is replaced with an oscillator SG2 variable in frequency and in
which a frequency control unit FC is added. Shown in FIG. 53 is a
frequency control signal fco supplied from the frequency control unit FC
to the oscillator SG2. Given to the frequency control unit FC are a
threshold judgment unit output cto and a synchronizing trigger signal st.
In FIG. 53, the oscillator SG2 is formed of a PLL synthesizer. As far as
the oscillator SG2 is highly stable in view of temperature and variable in
frequency, high-speed pulling-into-synchronism properties are not
particularly required. If the oscillator SG2 is formed with the frequency
fixed by frequency multiplication, there occurs an frequency error (1 ppm
to 50 ppm) due to crystal frequency precision. This means that, when the
frequency fc of the oscillator SG2 is set to 2484 Mhz and the channel
interval after down-conversion in frequency is set to 2.times..DELTA.f=500
kHz, an error of 10 ppm will result in frequency error of about 25 kHz.
FIG. 54 shows, in the form of a logarithm, the levels of spurious responses
that reception carrier frequency F(k) having frequency fb gives to
adjacent frequency bands. When the reception carrier frequency is
down-converted in frequency with a frequency error of 25 kHz, a spurious
response of about -25 dB is generated in the adjacent carrier frequencies
F(k-1) and F(k+1). According to the arrangement in FIG. 53, based on the
degree of difference in spectrum value, the PPL synthesizer forming the
oscillator SG2 is controlled in frequency using the frequency control
signal fco generated by the frequency control unit FC. Thus, the
oscillator frequencies fc of a plurality of digital communication
apparatus can be unified. This lowers the influence of spurious responses
to improve the carrier frequency detection precision.
FIG. 55 shows an example of the arrangement of the digital communication
apparatus according to the present invention. Shown at the transmitter T
in FIG. 55 are transmission data dt, a frequency selection unit 1SF, a
digital direct synthesizer 1DDS, an in-phase-axis mixer 1MI, a
quadrature-axis mixer 1MQ, a 90.degree. phase shifter 1PS, an adder 1ADD
and a power amplifier 1PA. Shown at the receiver R in FIG. 55 are a
band-pass filter 1BFR, a low noise amplifier 1LNA, a mixer 1MR, a low-pass
filter 1LFR, an automatic gain controller (AGC) 1AGC, an A/D converter
1AD, a discrete Fourier transform (DFT) operation unit 1DFT, a window
control unit 1WC, a level judgment unit 1DT and reception data dr. Also
shown in FIG. 55 are an antenna 1AT, an antenna switch 1SW, a first
oscillator 1SGA for generating a reference oscillating signal having
frequency fc.sub.-- a1, a second oscillator 1SGB for generating a
reference oscillating signal having frequency fc.sub.-- b1, a third
oscillator 1SGC for generating a reference oscillating signal having
frequency fc.sub.-- c1, a pseudorandom noise (PN) generator 1PN and a
selector 1SL.
In the digital communication apparatus in FIG. 55, there is used an MFSK
mode or a code multiplexing MFSK mode using M carrier frequencies per
sub-band, M being an integer not less than 4. The digital communication
apparatus in FIG. 55 is arranged such that when transmission data dt are
entered into the transmitter T, a transmission signal is supplied per time
slot and that when a reception signal is entered into the receiver R, all
the received frequencies are supplied per time slot. The antenna 1AT and
the antenna switch 1SW form a front end unit. The three oscillators 1SGA,
1SGB, 1SGC, the PN generator 1PN and the selector 1SL form a reference
oscillator unit for changing the local oscillating frequency according to
low-speed hopping of the desired sub-band. In the transmitter T, the
frequency selection unit 1SF determines the frequencies according to the
transmission data dt. The digital direct synthesizer LDDS generates, based
on the frequencies thus determined, base band signals of two series for
in-phase-axis components and quadrature-axis components. A modulation unit
composed of the in-phase-axis mixer 1MI, the quadrature-axis mixer 1MQ,
the 90.degree. phase shifter 1PS and the adder 1ADD, orthogonally
modulates local oscillating frequencies according to the two-series base
band signals. The power amplifier 1PA executes a signal amplification such
that the output of the modulation unit is supplied from the front end
unit. In the receiver R, the band-pass filter 1BFR limits in band a
reception signal entered from the front end unit. The low noise amplifier
1LNA amplifies, by predetermined gain, the signal limited in band. A
down-converter unit formed of the mixer 1MR converts in frequency an
output of the low noise amplifier 1LNA to a low frequency band using the
local oscillating frequencies. The low-pass filter 1LFR takes out, from an
output of the down-converter unit, a signal component for a 1/2 band width
of the sub-band. The AGC amplifier 1AGC amplifies an output of the
low-pass filter 1LFR up to the normalization level. The A/D converter 1AD
converts an output of the AGC amplifier 1AGC into a digital value. In the
DFT operation unit 1DFT, an output of the A/D converter 1AD is subjected
to a discrete Fourier transform (DFT). In the window control unit 1WC, the
DFT window is synchronously controlled based on an output of the DFT
operation unit 1DFT. The level judgment unit 1DT supplies reception
frequency data dr obtained by judging in level the output of the DFT
operation unit 1DFT.
The following description will discuss the operations of the respective
units in the arrangement in FIG. 55 in which M is equal to 8 and three
sub-bands are to be hopped at low speed.
In the frequency selection unit 1SF in the transmitter T, serial
transmission data dt are divided, for each period of one time slot T, into
blocks each having log.sub.2 M bits. In the digital direct synthesizer
LDDS, according to the signal from the window control unit 1WC, base band
signals BI(k) and BQ(k) (k=1, 2, . . . , M) for in-phase axis and
quadrature axis, are supplied, in synchronism with the time slot, for each
block according to the following equations (9) to (12):
BI(k=odd number)=cos (2.pi..times..DELTA.f.times.(2k-1).times.t)(9)
BQ(k=odd number)=sin (2.pi..times..DELTA.f.times.(2k-1).times.t)(10)
BI(k=even number)=cos (2.pi..times..DELTA.f.times.(2k-1).times.t)(11)
BQ(k=even number)=-sin (2.pi..times..DELTA.f.times.(2k-1).times.t)(12)
in which .DELTA.f means the frequency step width (.DELTA.f=1/T) and t means
time.
The PN generator 1PN generates a pattern for hopping a sub-band at low
speed. In the selector 1SL, there is selected, based on the pattern thus
generated, one of the frequencies of the three oscillators 1SGA, 1SGB,
1SGC, i.e., fc.sub.-- a1, fc.sub.-- b1, fc.sub.-- c1. It is now supposed
that the frequency fc is selected. The signal of frequency fc is
multiplied by the base band signal BI(k) in the mixer 1MI and also
multiplied by the base band signal BQ(k) in the mixer 1MQ after the signal
has passed through the 90.degree. phase shifter 1PS. The respective
products are added to each other in the adder 1ADD. Thus, an orthogonal
modulation signal W is obtained according to the following equation (13):
##EQU6##
The frequency interval between fc.sub.-- a1 and fc.sub.-- b1 and the
frequency interval between fc.sub.-- b1 and fc.sub.-- c1 are set to
M.times.4.times..DELTA.f such that the sub-bands do not overlap one
another. The frequencies of the sub-bands are maintained as orthogonal.
The orthogonal modulation signal W is amplified by the power amplifier 1PA
and then supplied from the antenna 1AT through the antenna switch 1SW.
FIG. 56 shows the arrangement of frequencies of the three sub-bands Sa1,
Sb1, Sc1, at certain time, used in the digital communication apparatus in
FIG. 55. A signal received from the antenna 1AT, is entered into the
receiver R through the antenna switch 1SW. In the receiver R, the
band-pass filter 1BFR takes out the signal components in the desired band
alone. The low noise amplifier 1LNA amplifies, by predetermined gain, the
signal components thus taken. In the mixer 1MR, the output of the low
noise amplifier 1LNA is down-converted to a base band frequency band
using, out of the frequencies (fc.sub.-- a1, fc.sub.-- b1, fc.sub.-- c1)
of the three oscillators 1SGA, 1SGB, 1SGC, the frequency which is
synchronized with the low-speed hopping pattern of the desired sub-band.
FIG. 57 shows the frequency arrangement obtained after the first sub-band
Sa1 has been down-converted using the frequency fc.sub.-- a1, FIG. 58
shows the frequency arrangement obtained after the second sub-band Sb1 has
been down-converted using the frequency fc.sub.-- b1 and FIG. 59 shows the
frequency arrangement obtained after the third sub-band Sc1 has been
down-converted using the frequency fc.sub.-- c1. In each of FIGS. 57 to
59, a broken line shows the frequency characteristics of the low-pass
filter 1LFR. In each of FIGS. 57 to 59, signal components not greater than
the frequency fc are turned back on the basis of the DC (0 Hz) point, but
are arranged in gaps between frequencies not less than fc with the
orthogonal relationship maintained. After down-conversion, a 1/2 band
width of the sub-band is taken out by the low-pass filter 1LFR, amplified
up to a predetermined level by the AGC amplifier 1AGC and then converted
into a digital value by the A/D converter 1AD. After the digital signal
level of frequency has been calculated by the DFT operation unit 1DFT and
the window control unit 1WC, the reception frequencies are determined by
the level judgment unit 1DT to obtain reception data dr.
Thus, according to the arrangement in FIG. 55, even in the environment
where simultaneous communications are conducted with a plurality of
sub-bands, frequencies in the sub-band around a specific frequency can be
detected using a low sampling-rate discrete Fourier transform capable of
processing a 1/2 band width of a sub-band. This can also be applied to the
case where the number of sub-bands is 2 or not less than 4.
FIG. 60 shows a modification of the arrangement in FIG. 55. In the
arrangement in FIG. 60, there is used a reference oscillation unit
composed of one oscillator 1SG, a phase locked loop circuit 1PLL and a PN
generator 1PN. This arrangement achieves a low-speed hopping having an
overlap of sub-bands. The operations of other units are the same as those
in FIG. 55. More specifically, the digital communication apparatus in FIG.
60 is arranged to use an MFSK mode or a code multiplexing MFSK mode, using
M consecutive carrier frequencies randomly selected per predetermined time
interval L with M set to an integer not less than 4. Here, the time
interval L is a value equal to the product of a period of one time slot T
(=1/.DELTA.f) and a positive integer.
FIG. 61 shows the arrangement of frequencies of two sub-bands Sa2, Sb2, at
certain time, used in the digital communication apparatus in FIG. 60 in
which M is equal to 8 and eight consecutive carrier frequencies are to be
randomly selected out of 16 carrier frequencies. The 16 carrier
frequencies are orthogonally disposed at frequency intervals of
4.times..DELTA.f.
FIG. 62 shows the frequency arrangement obtained after the first sub-band
Sa2 has been down-converted using the frequency fc.sub.-- a2, and FIG. 63
shows the frequency arrangement obtained after the second sub-band Sb2 has
been down-converted using the frequency fc.sub.-- b2. In each of FIGS. 62
to 63, a broken line shows the frequency characteristics of the low-pass
filter 1LFR.
According to the arrangement in FIG. 60, even in the environment where
simultaneous communications are conducted with a plurality of sub-bands,
frequencies in the sub-band around a desired frequency can be detected
using a low sampling-rate discrete Fourier transform capable of processing
a 1/2 band width of a sub-band. This can also be applied to the case where
the number of sub-bands is not less than 3.
FIG. 64 shows a modification of the receiver R in FIG. 55 with M equal to
16. In FIG. 64, the reference oscillation unit in FIG. 55 is replaced with
a single oscillator 7SG. That is, the number of the sub-bands is equal to
1. In FIG. 64, three antennas 7AT1, 7AT2, 7AT3 are disposed as spatially
separated from one another such that the fading influences are independent
from one another (without any correlation). The receiver R comprises
diversity branches 7DB1, 7DB2, 7DB3, a selector 7SLD, a frequency
detection unit 7PR having 16 processing units, a level judgment unit 7DT,
a window control unit 7WC and a timer 7TM. Also shown in FIG. 64 is a time
slot synchronizing signal 7wco. Each of the three diversity branches 7DB1,
7DB2, 7DB3 comprises a band-pass filter 1BFR, a low noise amplifier 1LNA,
a mixer 1MR, a low-pass filter 1LFR, an AGC amplifier 1AGC and an A/D
converter 1AD. These three diversity branches 7DB1, 7DB2, 7DB3
respectively down-convert in frequency the signals received from three
spatially separated points to low frequency bands, thus supplying
3-sequence base band signals.
FIG. 65 shows in detail an example of the arrangement of one operation unit
in FIG. 64. The operation unit 7PR(k) in FIG. 65 comprises a cosine-wave
memory 13CRM, a sine-wave memory 13SRM, multipliers 13CMX, 13SMX,
accumulators 13CAC, 13SAC, delay units 13CDL, 13SDL, latches 13CLT, 13SLT
and an absolute value operation unit 13ABS. According to the operation
unit 7PR(k) having the arrangement above-mentioned, there are achieved
both a correlation operation for the desired frequency out of the 16
frequencies and a signal intensity calculation using the result of the
correlation operation. That is, there are accumulated complex correlation
values for one time slot between the base band signal after A/D conversion
from the selector 7SLD and the cosine and sine waves of the desired
frequency. Based on the accumulated 2-sequence components, the signal
intensity is calculated by an operation of the absolute values of complex
numbers.
FIGS. 66A, 66B and 66C show the frequencies received by the three diversity
branches 7DB1, 7DB2, 7DB3 under the influence of fading. The broken line
in FIG. 66A shows the reception characteristics of the first diversity
branch 7DB1, the broken line in FIG. 66B shows the reception
characteristics of the second diversity branch 7DB2 and the broken line in
FIG. 66C shows the reception characteristics of the third diversity branch
7DB3. The frequency detection unit 7PR formed of 16 operation units,
calculates the frequency levels for the first diversity branch 7DB1. In
FIG. 66A, the first to eighth frequencies cannot be received due to fading
influence. Accordingly, through the selector 7SLD, the timer 7TM changes
the assignment of a diversity branch to operation units in each of which
no frequency has been detected in a determined period of time. That is,
the second diversity branch 7DB2 is assigned to the first to eighth
operation units. In FIG. 66B, the sixth to eighth frequencies can be
received by the second diversity branch 7DB2, but the first to fifth
frequencies cannot be received due to fading influence. Through the
selector 7SLD, the timer 7TM assigns the third diversity branch 7DB3 to
the first to fifth operation units after a subsequent predetermined period
of time. This enables the first to fifth frequencies to be received by the
third diversity branch 7DB3 as shown in FIG. 66C.
As discussed in the foregoing, the arrangement in FIG. 64 enables
frequencies to be received with no fading influence exerted to each
operation unit. When two or more diversity branches are disposed, similar
effects can be produced. When the oscillator 7SG is replaced with a
reference oscillation unit in FIG. 55 or 60, similar effects can also be
produced even with a plurality of sub-bands. When fading varies relatively
at high speed, the switching period of time of the timer 7TM may be
shortened.
FIG. 67 shows a digital communication apparatus to be used for a digital
communication system in which a plurality of digital communication
apparatus share a time slot and in which a half-duplex data communication
is made using an MFSK mode or a code multiplexing MFSK mode. This digital
communication apparatus is characterized in that a regenerative
synchronizing signal is generated for a synchronous control of the time
slot by a feedback control, in the reception mode, based on the detected
phase error and by a feedforward control, in the transmission mode, based
on the phase error stored immediately before the start of transmission.
Shown in FIG. 67 are an antenna switch 16SW, a DFT operation unit 16DFT, a
window control unit 16WC, a mode control signal 16md for
transmission/reception changeover, a DFT operation unit output 16dfto and
a regenerative synchronizing signal 16rsc. The antenna 1AT is commonly
used in both the transmission and reception modes through the antenna
switch 16SW, and used in time division according to the mode control
signal 16md. The regenerative synchronizing signal 16rsc from the window
control unit 16WC is used for controlling the digital direct synthesizer
1DDS in the transmitter T and the DFT operation unit 16DFT in the receiver
R such that the synthesizer 1DDS and the unit 16DFT are operated in
synchronism with the time slot.
FIG. 68 shows in detail an example of the arrangement of the window control
unit 16WC in FIG. 67. Shown in FIG. 68 are a timing detection unit 17TMD,
a phase error detection unit 17PED, a timer 17TM, a phase error memory
unit 17PEM, a crystal oscillator 17CSG, a phase error correction unit
17PEC and a time slot edge information 17edg.
When the mode control signal 16md is negated, the timing detection unit
17TMD takes out, from the DFT operation unit output 16dfto, the time slot
edge information 17edg which undergoes a momentary change (presenting a
jitter). The phase error detection unit 17PED detects a time-average phase
error between the current regenerative synchronizing signal 16rsc and the
timing detection unit output 17edg. In the phase error memory unit 17PEM,
the phase error thus detected is stored as rewritten at predetermined time
intervals. The timer 17TM controls the phase error correction unit 17PEC
such that the regenerative synchronizing signal 16rsc is feedbacked to the
phase error detection unit 17PED at predetermined time intervals on the
basis of an output of the crystal oscillator 17CSG.
FIG. 69 shows the timing chart of the operations of the window control unit
16WC when the mode control signal 16md is asserted. When the mode control
signal 16md is asserted from the LOW level to the HIGH level, the timer
17TM controls the phase error detection unit 17PED such that there is
calculated the latest offset phase error generated in a predetermined
period of time Ts after the phase error has been corrected. The phase
error thus calculated is stored as a reference phase error in the phase
error memory unit 17PEM, and then held in a unrewritable manner. In the
transmission mode, the timer 17TM causes the phase error correction unit
17PEC to feedforward-control, per period of time Ts, the correction of the
constant reference phase error. When the mode control signal 16md is
negated and the digital communication apparatus is returned to the
reception mode, the apparatus is returned to the feedback-control of the
regenerative synchronizing signal 16rsc.
As discussed in the foregoing, according to the arrangement in FIG. 67, it
is possible to maintain a network synchronization at the time when there
is made, using the common antenna, a code division multiple access (CDMA)
as done in an FH-MFSK mode in the same frequency band.
FIG. 70 shows an example of the arrangement of the digital communication
apparatus according to the present invention. In FIG. 70, the transmitter
T comprises a coding unit SO0, a convolutional coder SO1, an interleaver
SO2, an FH coder SO3, a switching unit SO4 and an M-ary independent signal
transmission unit SO5. The receiver R in FIG. 70 comprises a decoding unit
RO0, an M-ary independent signal reception unit RO1, an operational mode
control unit RO2, an FH decoder RO3, a switching unit RO4, a majority
decoder RO5, a deinterleaver RO6 and a Viterbi decoder RO7.
In the transmitter T, the convolutional coder SO1 supplies, to the
interleaver SO2, a convolutional code sequence according to an entered
information sequence. The convolutional coder SO1 and the interleaver SO2
give, to the input information sequence, invulnerability to random errors
and burst errors. An interleave sequence supplied by the interleaver SO2
is given to the FH coder SO3 and the switching unit SO4. The FH coder SO3
causes the interleave sequence to be coded using a multiplexing code such
that one word is extended to L words. Here, it is noted that M is an
integer not less than 2 and L is an integer not greater than M. An FH code
sequence supplied by the FH coder SO3 is given to the switching unit SO4.
The switching unit SO4 selects, as a transmission sequence, the FH code
sequence when the switching signal supplied from the operational mode
control unit RO2 so instructs as to execute an FH coding, and the
interleave sequence when the switching signal instructs otherwise. The
M-ary independent signal transmission unit SO5 supplies, per time slot, a
transmission signal containing, out of mutually independent M frequency
components, one frequency component corresponding to the transmission
sequence (M-ary sequence) supplied from the switching unit SO4.
In the receiver R, the M-ary independent signal reception unit RO1
supplies, as a threshold judgment pattern, the results obtained by
conducting a threshold judgment on the intensity values of the M frequency
components of the reception signal. The threshold judgment pattern is
given to the operational mode control unit RO2, the FH decoder RO3 and the
switching unit RO4. The operational mode control unit RO2 judges the
multiplicity based on the threshold judgment pattern, and supplies a
switching signal for instructing to execute an FH coding/decoding when the
multiplicity is not less than 2, and a switching signal for instructing
not to execute the FH coding/decoding when the multiplicity is equal to 1.
The FH decoder RO3 decodes the threshold judgment pattern using a
multiplexing code, and supplies the decoding result as an FH decoding
pattern. The switching unit RO4 selects the FH decoding pattern when the
switching signal instructs to execute the FH decoding, and the threshold
judgment pattern when the switching signal instructs not to execute the FH
decoding. The majority decoder RO5 executes a majority judgment of each of
the bits of the pattern selected by the switching unit RO4, and determines
one word out of M different candidate words, which is then supplied as a
majority decoding sequence. The deinterleaver RO6 supplies, as a
deinterleave sequence, the result of interleave release of the majority
decoding sequence. The Viterbi decoder RO7 supplies, as an information
sequence, the result of error correction of the deinterleave sequence.
FIG. 71 shows in detail an example of the arrangement of the convolutional
coder SO1 in FIG. 70 in which the coding rate is set to 1/2 and the
constraint length is set to 7. In FIG. 71, there are disposed delay units
SCO1 to SCO6 and adders SCO7 to SC14 for calculating an exclusive OR. The
convolutional coder SO1 codes a given information sequence by convoluting,
using the adders SCO7 to SC14, past information sequences stored in the
delay units SCO1 to SCO6, and then supplies the coded result as a
convolutional code sequence.
FIG. 72A and FIG. 72B respectively show in detail examples of the
arrangements of the interleaver SO2 and the deinterleaver RO6 in FIG. 70.
Shown in FIG. 72 are serial-to-parallel converters SI1 to SI2,
parallel-to-serial converters RI1 to RI2, shift registers SI3, RI5 each
having a length of B, shift registers SI4, RI4 having a length of 2B and
shift registers SI5, RI3 having a length of 3B. It is noted that B is a
positive integer. In the interleaver SO2, a convolutional code sequence
entered in 2-bit parallel is converted, by the serial-to-parallel
converters SI1, SI2, into a 4-bit parallel sequence, which passes through
the shift registers SI3, SI4, SI5 respectively having different lengths.
This causes the respective bits to be arranged as dispersed in a time
direction, thus forming an interleave sequence. In the deinterleaver RO6,
a given majority decoding sequence is passed through the shift registers
RI3, RI4, RI5 disposed in the reverse order with respect to the shift
registers in the interleaver SO2 such that the respective bits dispersed
in the time direction are returned back, and two-bit data are converted
into serial data by the parallel-to-serial converters RI1, RI2, thus
forming a 2-bit parallel deinterleave sequence. The value of B is set such
that errors in the transmission line are sufficiently dispersed in the
time direction. As the set value of B is greater, the invulnerability to
burst error is stronger.
FIG. 73A and FIG. 73B respectively show in detail examples of the
arrangements of the FH coder SO3 and the FH decoder RO3 in FIG. 70. Shown
in FIG. 70 are an adder SF1 modulo M, a subtracter RF1 modulo M,
multiplexing code generators SF2, RF2 and a majority logic judgment unit
RF3. In the FH coder SO3 in FIG. 73A, the adder SF1 executes addition
modulo M on the interleave sequence and the multiplexing code supplied
from the multiplexing code generator SF2, and the addition result is
supplied as the FH code sequence. This causes one word of the interleave
sequence to be divided into L time elements such that L time elements are
randomly spread in the level direction. In the FH decoder RO3 in FIG. 73B,
the subtracter RF1 subtracts (inversely spreads) the multiplexing code
supplied from the multiplexing code generator RF2, from each of the level
values of the threshold judgment pattern, and supplies the subtraction
results to the majority logic judgment unit RF3. Based on the output of
the subtracter RF1, the majority logic judgment unit RF3 judges the level
value containing the most numerous time elements as a proper level value,
and then supplies the judgment result as the FH decoding pattern.
FIGS. 74A, 74B, 74C respectively show examples of the interleave sequence
matrix, multiplexing code matrix and FH code sequence matrix in the FH
coder SO3 in FIG. 73A. FIGS. 75A, 75B, 75C, 75D respectively show examples
of the threshold judgment pattern matrix, multiplexing code matrix,
judgment matrix and FH decoding pattern matrix in the FH decoder RO3 in
FIG. 73B. The judgment matrix in FIG. 75C represents an output of the
subtracter RF1. In FIGS. 74 and 75, crosses show the level values per word
of each sequence when M is equal to 16 and L is equal to 8, and circles
show the level values of an undesired sequence resulting from other user
in the FH decoder RO3. The threshold judgment pattern matrix in FIG. 75A
contains both the level values of the desired sequence (crosses) and the
level values of the undesired sequence (circles). However, the inverse
spread causes the desired sequence to be arranged in a line of a specific
level and causes the undesired sequence to be randomly dispersed in the
level direction as shown in the judgment matrix in FIG. 75C. Accordingly,
there is obtained, by the majority logic judgment unit RF3, the FH
decoding pattern containing only the desired sequence as shown in FIG.
75D. In the threshold judgment pattern, however, a level value which is
not to be present, is generated due to noise or a spurious response, and a
level value which is to be present, is erased due to miss detection. In
this connection, there are instances where the result of majority judgment
is erroneous or where a majority judgment itself cannot be made (where
there are present a plurality of level values each of which contains the
most numerous elements). When an error takes place, the Viterbi decoder
RO7 in FIG. 70 executes an error correction. When the majority judgment
itself cannot be made, all the level values each containing the most
numerous elements are supplied and a majority judgment is made on each bit
by the majority decoder RO5 in FIG. 70.
FIGS. 76A and 76B respectively show in detail examples (M=16) of the
arrangements of the M-ary independent signal transmission unit SO5 and the
M-ary independent signal reception unit RO1 in FIG. 70. Shown in FIGS. 76A
and 76B are a tone generator SM1, an up-converter SM2, band pass filters
RM01 to RM16 respectively having center frequencies f1 to f16, intensity
detectors RM17 to RM32 and threshold judgment units RM33 to RM48. The
M-ary independent signal transmission unit SO5 executes an MFSK modulation
on a hexadecimal transmission sequence, and the M-ary independent signal
reception unit RO1 executes an MFSK demodulation on a reception signal. In
the Mary independent signal transmission unit SO5, the tone generator SM1
generates frequency tones for the word values of a given transmission
sequence, and the up-converter SM2 pulls the frequency tones up to the
desired band, which is then supplied as a transmission signal. In the
M-ary independent signal reception unit RO1, the band pass filters RM01 to
RM16 take out frequency components f1 to f16 from the reception signal.
The intensity detectors RM17 to RM32 detect the signal intensities of the
frequency components. The threshold judgment units RM33 to RM48 make a
threshold judgment on the signal intensities and supply the results as a
2.sup.16 -ary threshold judgment pattern. The value of each of the bits of
the threshold judgment pattern is set to 1 when the signal intensity
exceeds the threshold value, and to 0 otherwise. The threshold judgment
pattern contains the desired sequence transmitted from the desired user
and an undesired sequence transmitted from other user.
FIG. 77 shows in detail an example of the arrangement of the operational
mode control unit RO2 in FIG. 70. Shown in FIG. 77 are a multiplicity
judgment logic RW1, a shift register RW2, an adder RW3 and a changeover
judgment unit RW4. The operational mode control unit RO2 in FIG. 77 makes
a judgment based on the threshold judgment pattern whether the
multiplicity is singular or plural, and supplies a binary switching signal
to be entered into the two switching units SO4, RO4 in FIG. 70. More
specifically, the multiplicity judgment logic RW1 supplies a logical value
of "0" when the Hamming weight of the threshold judgment pattern (the
number of bits each presenting a value 1) is equal to 0 or 1, and a
logical value of "1" when the Hamming weight is not less than 2. The
logical value "0" represents that the multiplicity is singular, while the
logical value "1" represents the multiplicity is plural. However, there
are instances where the multiplicity here displayed is not accurate due to
noise or a spurious response. Such an error often appears in the form of
subtle variations. To remove such variations to enhance the reliability of
operational mode control, the operational mode control unit RO2 in FIG. 77
is so arranged as to adopt, out of two (singular and plural) judgement
values of multiplicity, the judgment value of multiplicity for which the
number of judgment times is larger. In this connection, the shift register
RW2 successively delays the logical value supplied from the multiplicity
judgment logic RW1 and supplies, to the adder RW3, the result as an n-bit
parallel sequence. It is noted that n is an odd number not less than 3.
The adder RW3 calculates the Hamming weight of the n-bit parallel
sequence. The changeover judgment unit RW4 judges which is larger, the
result of the adder RW3 or the integer n/2. Then, the changeover judgment
unit RW4 supplies, as a switching signal, the logical value "0" when the
logical value "0" appears more often than the logical value "1", in the n
logical values obtained through the past n judgments, and the logical
value "1" when the logical value "1" appears more often.
FIG. 78 shows in detail an example of the arrangement (M=16) of the
majority decoder RO5 in FIG. 70. Shown in FIG. 78 are adders RC01 to RC08
and comparators RC09 to RC12. In the comparators RC09 to RC12, each of x
and y refers to a 4-bit (total 8-bit) comparator input, and each z refers
to a 1-bit comparator output. In FIG. 78, each of the numerals put to the
16 lines extending from the switching unit RO4 represents, in a binary
notation, the word value of the corresponding line. In an ideal
environment, a detection value of "1" indicative of a reception word
appears only on one line out of the 16 lines. Actually, however, the
detection value "1" appears on a plurality of lines under the influence of
noise or spurious responses. In this connection, the Hamming weight of an
input (the number of appearances of value "1" in the 8-bit parallel
sequence) is calculated. The calculated Hamming weight has a value of any
of 0 to 8, and represents the probability that each of the bits forming
one word is equal to 0 or 1. More specifically, each of the four
comparators RC09 to RC12 judges which input is larger out of the inputs
from two corresponding adders, and supplies 0 as z when x is smaller than
y, and 1 as z when x is not less than y. The outputs of the four
comparators RC09 to RC12 form a 4-bit majority decoding sequence. However,
when x is equal to y, the possibility of each of the bits forming one word
being equal to 0, is equal to the possibility of each bit being equal to
1. Therefore, it cannot be said which probability is higher. That is, the
bit for which x is equal to y, is an undeterminable bit. However, a bit
which is even an undeterminable bit, must be determined as 0 or 1. In the
majority decoder RO5 in FIG. 78, provision is made such that z is equal to
1 when x is equal to y. However, when the data equilibrium is established
(the probability of 0 appearing is equal to the probability of 1
appearing), z may be equal to 0 with the same performance provided.
The foregoing has discussed in detail examples of the arrangements of the
respective components of the digital communication apparatus in FIG. 70.
According to the arrangement in FIG. 70, when the convolutional coder SO1
and the interleaver SO2 are used as combined with each other in the
transmitter T, transmission data are randomized, thus producing a
transmission signal having an even frequency distribution. In the receiver
R, the majority decoder RO5 makes a majority judgment on each of the bits
forming a word, thereby to determine the maximum likelihood word. This
reduces errors which result from an undeterminable bit. It is noted that
the values of M and L are not limited to the examples above-mentioned
(M=16, L=8). Further, the coding rate and the constraint length in the
convolutional coder SO1 are not limited to the values in the examples
above-mentioned. Each of the adder SF1 and the subtracter RF1 may be
replaced with a circuit for calculating an exclusive OR per bit. In such a
case, the adder and the subtracter have the same circuit arrangement.
FIG. 79 shows a modification of the arrangement in FIG. 70. In FIG. 79, a
burst signal component removal circuit R08 is disposed downstream of the
M-ary independent signal reception unit RO1.
FIG. 80 shows in detail an example of the arrangement of the burst signal
component removal circuit R08 (M=16). Shown in FIG. 80 are a burst
detection circuit RB01, a burst removal circuit RB02, 16 bits BI1 to BI16
forming a threshold judgment pattern and 16 bits BO0 to BO16 forming a
burst removal pattern. The threshold judgment pattern BI1 to BI16
corresponds to 16 carrier frequencies. Each bit presents a value of "1"
when the corresponding carrier frequency is received, and a value of "0"
when the corresponding carrier frequency is not received. The burst
detection circuit RBO1 judges whether or not a value of "1" appears in the
form of a burst for the threshold judgment pattern bits BI1 to BI16, and
supplies the results as a burst judgment pattern. The burst removal
circuit RBO2 regards, out of the bits of the threshold judgment pattern,
the bit judged as a burst, as a false detection bit resulting from a
single carrier jamming wave, and then changes the value of such a bit to
"0" for invalidating the same. As an exception, however, when all the bits
not judged as bursts are equal to 0, the bits are not invalidated and the
threshold judgment pattern BI1 to BI16 is used, as it is, as a burst
removal pattern BO1 to BO16.
FIG. 81 shows in detail an example of the arrangement of each of the 16
burst detection units forming the burst detection circuit RBO1 in FIG. 80.
Shown in the burst detection unit RBO1(k) in FIG. 81 are a shift register
RBO3, an adder RBO4, a burst judgment unit RBO5, each bit BI of the
threshold judgment pattern and each bit J of the burst judgment pattern.
The shift register RBO3 successively delays the bits BI forming the
threshold judgment pattern and supplies, to the adder RBO4, the results as
a p-bit parallel sequence. Here, p is an integer not less than 2. The
adder RBO4 calculates the Hamming weight of the p-bit parallel sequence.
The burst judgment unit RBO5 judges which is larger, the result of the
adder RBO4 or an integer q. Thus, the burst properties of the value "1"
are judged. Here, q is an integer not less than 0 and not greater than p.
More specifically, j is equal to 1 when the adder output is larger than q,
and j is equal to 0 when the adder output is not greater than q.
FIG. 82 shows in detail an example of the arrangement of the burst removal
circuit RBO2 in FIG. 80. Shown in FIG. 82 are burst removal logic units
RBO6, an OR circuit RBO7, 16 bits BI1 to BI16 forming the threshold
judgment pattern, 16 bits J1 to J16 forming the burst judgment pattern, 16
bits BO1 to BO16 forming the burst removal pattern, 16 bits NB1 to NB16
forming a non-burst detection signal and a burst bit invalidating signal
DEL.
FIG. 83 shows in detail an example of the arrangement of each of the 16
burst removal logic units RBO6. Shown in FIG. 83 are inverter circuits
RB11, RB12, AND circuits RB13, B14, RB15, each bit BI of the threshold
judgment pattern, each bit J of the burst judgment pattern, the burst bit
invalidating signal DEL, each bit BO of the burst removal pattern and each
bit NB of the non-burst detection signal.
Each burst removal logic unit RBO6 in FIG. 82 generates a non-burst
detection signal NB based on the threshold judgment pattern BI and the
burst judgment pattern J. NB is equal to BI when J is equal to 0, and NB
is equal to 0 when J is equal to 1. Then, the logical OR of the 16 bits
NB1 to NB16 forming the non-burst detection signal, is supplied to each
burst removal logic unit RBO6 as the burst bit invalidating signal DEL.
More specifically, the burst bit invalidating signal DEL instructs to each
burst removal logic unit RBO6 that, when at least one of the values of the
non-burst detection signal bits NB1 to NB16 is equal to 1, the burst bit
is invalidated. Each burst removal logic unit RBO6 causes BO to be equal
to 0 when DEL is equal to 1 and J is equal to 1, and causes BO to be equal
to BI otherwise.
Thus, according to the arrangement in FIG. 79, the burst signal component
removal circuit RO8 removes consecutive constant signal components, thus
lowering the influence of a jamming wave in a specific frequency band.
FIG. 84 shows a further modification of the arrangement in FIG. 70. In the
decoding unit RO0 in FIG. 84, a puncture signal generator RO9 is disposed
downstream of the switching unit RO4, a deinterleaver R10 is disposed
downstream of the puncture signal generator RO9, and a Viterbi decoder R11
is arranged to process a puncture signal input.
FIG. 85 shows in detail an example of the arrangement of the puncture
signal generator RO9 (M=16). Shown in FIG. 85 are adders RCO1 to RCO8 and
comparators RC13 to RC16. In the comparators RC13 to RC16, each of x and y
refers to a 4-bit comparator input, and each eq refers to a 1-bit
comparator output. Except for the operations of the comparators RC13 to
RC16, the puncture signal generator RO9 is the same as the majority
decoder RO5 shown in FIG. 78. Each of the four comparators RC13 to RC16
compares the inputs from the two corresponding adders with each other, and
supplies 0 as eq when x is not equal to y, and 1 as eq when x is equal to
y. The outputs of the four comparators RC13 to RC16 form a 4-bit puncture
signal. That is, the puncture signal generator RO9 is arranged to display
an undeterminable bit. More specifically, when a certain bit of a puncture
signal is equal to 1, the corresponding bit of an output of the majority
decoder RO5 is an undeterminable bit.
As shown in FIG. 84, to maintain the corresponding relationship with the
sequence obtained by the majority decoder RO5, the puncture signal is
processed by the deinterleaver R10 having the inside arrangement identical
with that of the deinterleaver RO6, and is then entered into the Viterbi
decoder R11 as a deinterleave-puncture signal. While a bit designated by
the deinterleave-puncture signal is handled as an erasure bit, the Viterbi
decoder R11 executes a Viterbi decoding on the sequence supplied from the
deinterleaver RO6. Thus, when it is difficult to judge whether an output
bit of the majority decoder RO5 is "0" or "1", such an output bit is
handled as erasure without any judgment forcibly made thereon. Thus, a
more accurate error correction can be made. Since the Viterbi decoding of
a punctured code is an already established technique, the detailed
description thereof is here omitted.
Thus, according to the arrangement in FIG. 84, the undeterminable bit
designated by the puncture signal from the puncture signal generator RO9,
is regarded as an erasure bit in the Viterbi decoder R11, enabling an
error correction to be made more efficiently.
FIG. 86 shows a further modification of the arrangement in FIG. 70. In the
decoder RO0 in FIG. 86, a multi-level decoder R12, a multi-level
deinterleaver R13 and a soft decision Viterbi decoder R14 are disposed
downstream of the switching unit RO4.
FIG. 87 shows in detail an example of the arrangement of the multi-level
decoder R12 (M=16). Shown in FIG. 87 are adders RCO1 to RCO8 and
comparators RC17 to RC20. In the comparators RC17 to RC20, each of x and y
refers to a 4-bit comparator input, and each mz refers to a 3-bit
comparator output. Except for the operations of the comparators RC17 to
RC20, the multi-level decoder R12 is the same as the majority decoder RO5
shown in FIG. 78. Each of the four comparators RC17 to RC20 executes an
operation of the following equation (14):
mz=x/(x+y) (14)
The outputs of the four comparators RC17 to RC20 form a multi-level
decoding sequence, more specifically, a 12-bit 3-level decoding sequence.
As compared with a binary judgement made in each of the comparators of the
majority decoder RO5 in FIG. 78, each of the comparators of the
multi-level decoder R12 in FIG. 87 makes an octal decision (soft
decision). Accordingly, the possibility of a decoding bit being equal to
"0" or "1", is displayed more finely. As the value of mz is larger, the
possibility of a decoding bit being equal to "1" is greater, and as the
value of mz is smaller, the possibility of the decoding bit being equal to
"0" is greater.
As shown in FIG. 86, a 3-level decoding sequence is supplied to the
multi-level deinterleaver R13. The multi-level deinterleaver R13 is formed
of three deinterleavers each having the same inside arrangement as that of
the deinterleaver RO6 in FIG. 72B. This multi-level deinterleaver R13
supplies a 6-bit 3-level deinterleave sequence to the soft decision
Viterbi decoder R14. The soft decision Viterbi decoder R14 executes a soft
decision decoding of the 3-level deinterleave sequence. This soft decision
Viterbi decoder R14 has an error-correcting capability higher than that of
the Viterbi decoder RO7 in FIG. 70. Since the soft decision Viterbi
decoding is a known technique, the detailed description thereof is here
omitted.
Thus, according to the arrangement in FIG. 86, a soft decision Viterbi
decoding is executed on that sequence from the multi-level decoder R12, on
which a soft decision has been made in multiple levels, thus achieving a
more efficient error correction. It is noted that the number of soft
decision levels in the multi-level decoder R12 is not limited to the
numeral in the above-mentioned example, or 3.
FIGS. 88A and 88B respectively show modifications of the arrangements in
FIGS. 76A and 76B (M=16). The M-ary independent signal transmission unit
SO5 in FIG. 88A comprises PN-sequence generators SP1 to SP16 for
respectively generating different pseudorandom noise (PN) sequences PN1 to
PN16, and a switch SP17. The switch SP17 selects, out of the PN sequences
generated by the PN- sequence generators SP1 to SP16, a sequence
corresponding to the value of a transmission sequence, and supplies the
sequence thus selected as a transmission signal. The M-ary independent
signal reception unit RO1 in FIG. 88B comprises PN-sequence generators RP1
to RP16 for respectively generating different PN sequences PN1 to PN16 and
correlation units RP17 to RP32. Each of the correlation units RP17 to RP32
calculates the correlation value between the received signal and the
corresponding PN sequence out of the 16 PN-sequences PN1 to PN16, and
supplies "1" when the correlation value thus calculated exceeds a
predetermined threshold value, and "0" when the correlation value does not
exceed the predetermined threshold value. The threshold value is so set as
to be smaller than the self-correlation value of the corresponding
PN-sequence and larger than the mutual correlation value with respect to
each of other PN sequences.
According to the arrangements in FIGS. 88A and 88B, a coder and a decoder
each of the direct spread (DS) type can be achieved. As the correlation
units RP17 to RP32, SAW convolvers may be used.
FIG. 89 shows a further modification of the arrangement in FIG. 70. In FIG.
89, the two switching units SO4, RO4 in FIG. 70 are omitted. Shown in FIG.
89 are an FH coder SO6, an operational mode control unit R15 and an FH
decoder R16. These circuit blocks are different in inside arrangement and
operation from the corresponding circuit blocks SO3, RO2, RO3 in FIG. 70.
A switching signal from the operational mode control unit R15 is supplied
to the FH coder SO6 and the FH decoder R16. Other arrangements are the
same as those in FIG. 70.
FIGS. 90A and 90B show in detail the arrangements of the FH coder SO6 and
the FH decoder R16 in FIG. 89. Shown in FIGS. 90A and 90B are an adder SF1
modulo M, a subtracter RF1 modulo M, code-length variable multiplexing
code generators SF3, RF4 and a majority logic judgment unit RF3. Unlike in
the FH coder SO3 and the FH decoder RO3 in FIGS. 73A and 73B, the
multiplexing code generators SF3, RF4 are arranged such that the length L
of a multiplexing code is variable according to a switching signal.
FIG. 91 shows in detail an example of the arrangement of the operational
mode control unit R15 in FIG. 89 (M=16). Shown in FIG. 91 are a
multiplicity judgment logic RW11, delay units RW12 to RW23, adders RW24 to
RW27 and a maximum value judgment logic RW28. It is now supposed that the
largest multiplicity is 4 and the number of operational modes is 4. The
operational mode control unit R15 in FIG. 91 calculates the multiplicity
based on a threshold judgment pattern and supplies a switching signal
(quaternary value) for switching the operational mode of each of the FH
coder SO6 and the FH decoder R16 to the operational mode corresponding to
the multiplicity. An output of the multiplicity judgment logic RW11 has
four bits y0 to y3.
FIG. 92 shows the relationships between the input and output of the
multiplicity judgment logic RW11. First, the multiplicity judgment logic
RW11 calculates, out of the 16 bits forming the threshold judgment
pattern, the number of bits represented by "1", and then judges the
multiplicity based on the bit number. In FIG. 92, only one bit out of the
four output bits y0 to y3 is always represented by "1". That is, the
multiplicity judgment logic RW11 shows the multiplicity based on the
output bits represented by "1". However, there are instances where the
multiplicity thus shown is not accurate under the influence of noise or a
spurious response. Such an error often appears in the form of subtle
variations. To remove such variations to enhance the reliability of
operational mode control, the operational mode control unit R15 in FIG. 91
is arranged to select the multiplicity which has been judged at the most
numerous frequency in a predetermined period of time. In this connection,
the 4 output bits y0 to y3 of the multiplicity judgment logic RW11 are
supplied, together with the past bits already entered into the delay units
RW12 to RW23, to the corresponding adders RW24 to RW27. The adders RW24 to
RW27 calculate the Hamming weights of the inputs, and display the number
of times of judgment made on each of the four different multiplicity
values. The maximum value judgment logic RW28 selects the multiplicity
which has been judged at the most numerous frequency, and then supplies a
switching signal of the operational mode corresponding to the multiplicity
thus selected.
Thus, according to the arrangement in FIG. 89, the operational mode control
unit R15 selects the multiplicity which has been judged at the most
numerous frequency, and then supplies a switching signal corresponding to
the multiplicity thus selected. This lowers the occurrence of errors about
the operational mode, thereby to achieve a highly reliable operational
mode control. Further, the operational mode can be switched finely
according to multiplicity, thus making a more efficient data transmission.
The largest multiplicity is selected from integers which are not less than
2 and not greater than M. The number of operational modes is selected from
integers which are not less than 2 and not greater than the largest
multiplicity. Each of the adder SF1 and the subtracter RF1 may be replaced
with a circuit for calculating an exclusive OR per bit. In this case, the
adder and the subtracter are the same in circuit arrangement.
FIG. 93 shows the arrangement of an asynchronous digital communication
system of the FH-MFSK mode of prior art. Shown in FIG. 93 are a
transmitter 20, a receiver 21, a transmission data input terminal 10, a
frequency hopping (FH) code generator 11 and a frequency synthesizer 12.
According to transmission data, the FH code generator (FH coder) 11
generates hopping codes, based on which the frequency synthesizer 12 hops
carrier frequencies. The FH code generator 11 utilizes a Reed-Solomon
code. The following shows one set of Reed-Solomon code vectors having
three chips based on a 4-element Galois field:
##EQU7##
Here, the vector components show the Nos. of carrier frequencies disposed
on a frequency band. The number of the components forming each vector
represents the number of chips for one hopping cycle.
Generally, when one of the primitive elements of a Q-element Galois field
is defined as .alpha., the spread code vector .alpha. of L components is
defined by the following equation (15):
.alpha.=(1, .alpha., .alpha..sup.2, .alpha..sup.L-1) (15)
wherein L means the number of chips per hopping cycle and L is smaller than
Q. When the user identification No. is defined as i, the data value is
defined as x and a unit vector of L components is defined as e=(1, 1, . .
. , 1), a hopping code vector y.sub.i (x) composed of L components is
calculated by the following equation (16):
y.sub.i (x)=x.multidot. .alpha.+i.multidot. e (16)
The user identification No. i, the data value x and the components of the
hopping code vector y.sub.i (x) are elements of the Q-element Galois
field. The operation represented by the equation (16) is an operation on
the Q-element Galois field. When Q is equal to 4(=2.sup.2) and L is equal
to 3, the equation (16) is modified as shown in the following equation
(17):
y.sub.i (x)=x.multidot.(1,2,3)+i.multidot.(1,1,1) (17)
FIGS. 94A and 94B respectively show the definitions of Galois addition and
Galois multiplication used in the FH code generator 11 in FIG. 93. For
example, the hopping code vector y.sub.2 (1) at the time when a user
having an identification No. i=2 transmits a data value x=1, is calculated
as shown in the following equation (18):
##EQU8##
FIG. 95 is a table showing hopping code vector values calculated in the
manner above-mentioned. Here, the data value x is equal to 0, 1, 2, or 3.
That is, when the number of values that the data can present, is defined
as M, M is equal to 4(=2.sup.2) and the relationship between M and the
number Q of the elements of the Galois field, is as follows:
Q=M
One set of hopping code vector shown in FIG. 95 is identical with one set
of a Reed-Solomon code vector having three chips based on the 4-element
Galois field mentioned earlier, and has an excellent feature that mutual
interference among users is very small in an asynchronous code multiple
communication system. As shown in FIG. 95, however, the hopping code
vector at the time when the user having an identification No. i=0
transmits a data value x=0, has three components each having the same
value. The hopping code vector at the time when other user transmits a
data value x=0, also has three components each having the same value. This
is a phenomenon taken place not only in the case where Q is equal to 4,
but also in each case where the equation (16) is adopted. In this system,
therefore, when a data value x=0, a signal having predetermined carrier
frequencies is transmitted. Thus, this system is susceptible to an
influence of frequency-selective fading.
FIGS. 96A and 96B respectively show time/frequency matrices in the
transmitter 20 and the receiver 21 in FIG. 93 under the influence of
frequency-selective fading. It is now supposed that a strong fading occurs
at frequency fd. Circles show predetermined carrier frequencies of a
transmission signal, while crosses show carrier frequencies for which a
miss detection has occurred due to fading. As shown in FIG. 96, there are
instances where a miss detection occurs in all the carrier frequencies at
the worst case.
To lower such an influence of frequency-selective fading, the present
invention is arranged such that a Q-element Galois field (Q is larger than
the number M of values that a data can present) is adopted, that a data
value x is previously converted into a non-zero code w and that a hopping
code vector y is calculated based on the code w, thus enhancing the
randomized property of frequency hopping codes.
The following description will discuss an example where M is equal to 4
(=2.sup.2), Q is equal to 5 and L is equal to 3. Using a 1:1 function f,
the data value x (0.ltoreq.x.ltoreq.3) is converted into a non-zero code w
(1.ltoreq.w.ltoreq.4). As an example of such a function, a function
f.sub.0 is defined using the following equation (19):
f.sub.0 (x)=x+1 (19)
Using the data value x, the code w is expressed as shown in the following
equation (20):
w=f.sub.0 (x) (20)
Then, the data value x in the equation (16) is replaced with the code w in
the equation (20). Then, the following equation (21) is obtained:
y.sub.i (w)=w.multidot. .alpha.+i.multidot. e (21)
Since .alpha. is equal to (1, 2.sup.1, 2.sup.2) and e is equal to
(1,1,1), the equation (21) is modified to the following equation (22):
##EQU9##
When the equation (22) is rewritten using the data value x, the following
equation (23) is obtained:
y.sub.i (x)=f.sub.0 (x).multidot.(1,2,4)+i.multidot.(1,1,1).multidot.(23)
FIGS. 97A and 97B respectively show the definitions of Galois addition and
Galois multiplication in the equations (22), (23). For example, the
hopping code vector y.sub.2 (1) at the time when a user having an
identification No. i=2 transmits a data value x=1, is calculated as shown
in the following equation (24):
##EQU10##
FIG. 98 shows a list of hopping code vectors calculated in the manner
above-mentioned. As shown in FIG. 98, a hopping code vector formed of
three components each having the same value, is never generated when any
user having any identification No. transmits any data value. This means
that the randomized property of a frequency hopping code is enhanced.
FIGS. 99A and 99B respectively show time/frequency matrices in a
transmitter and a receiver when there is adopted a frequency hopping code
enhanced in randomized property as above-mentioned. It is now supposed
that a strong fading occurs at frequency fd likewise in FIGS. 96A and 96B.
Since the carrier frequencies are dispersed, the number of carrier
frequencies for which a miss detection occurs, is advantageously
relatively small as shown in FIG. 99B.
FIG. 100 shows an example of the arrangement of an FH code generator (FH
coder) 400 in the digital communication apparatus according to the present
invention. The FH code generator 400 in FIG. 100 comprises a chip counter
40, a data converter 41, a spread code generator 42, a multiplier 43, an
adder 44, a data value x input terminal 401, a user identification No. i
input terminal 402 and a hopping code y output terminal 403. It is now
supposed that M is equal to 16 (=2.sup.4), Q is equal to 17 and L is equal
to 8. As shown in FIG. 101, the chip counter 40 executes a counting
operation modulo L=8 and supplies a counting value c to the data converter
41 and the spread code generator 42. Each time the counting value c is
equal to 0, the data converter 41 converts a data value x
(0.ltoreq.x.ltoreq.15) supplied from the input terminal 401, into a
non-zero code w (1.ltoreq.w.ltoreq.16). FIG. 102 shows a rule of
conversion from the data value x into the code w. As shown in FIG. 103,
the spread code generator 42 supplies, as a spread code pp, the primitive
element to the c-th power, that is .alpha..sup.c, which is corresponding
to the counting value c. Since L is equal to 8 in this example, it is
enough that the spread code generator 42 supplies a spread code pp=1, 2, .
. . , 11 corresponding to the counting value c=0, 1, . . . , 7. The
multiplier 43 executes Galois multiplication modulo Q=17 between the code
w obtained by the data converter 41 and the spread code pp supplied from
the spread code generator 42, and supplies the result mo to the adder 44.
The adder 44 executes Galois addition modulo Q=17 between the
multiplication result mo obtained by the multiplier 43 and the user
identification No. i supplied from the input terminal 402, and supplies,
through the output terminal 403, the addition result as a hopping code y.
This hopping code y is supplied to a frequency synthesizer having a choice
of at least 17 carrier frequencies.
According to the arrangement in FIG. 100, since the randomized property of
a frequency hopping code is enhanced, the influence of frequency-selective
fading can be reduced. The data converter 41 may adopt other conversion
rule.
FIG. 104 shows a modification of the arrangement in FIG. 100. An FH code
generator 600 in FIG. 104 comprises a chip counter 60, a read-only memory
(ROM) 61, a data value x input terminal 601 and a hopping code y output
terminal 602. The ROM 61 contains hopping codes previously calculated in
the manner above-mentioned and supplies a hopping code according to the
data value x and the counting value c. This arrangement eliminates the
need for calculation of a hopping code, thus achieving a high-speed
process.
FIG. 105 shows a further modification of the arrangement in FIG. 100. An FH
code generator 700 in FIG. 105 has means for calculating a binary judgment
vector formed of L components which shows whether or not each of the
components of an acquired hopping code vector y is contained in a list F
consisting of M different elements out of the Q elements of a Galois
field. Shown in FIG. 105 are a chip counter 70, a data converter 71, a
spread code generator 72, a multiplier 73, an adder 74, an FH code
judgment unit 75, a data value x input terminal 701, a user identification
No. i input terminal 702, a hopping code y output terminal 703 and a
binary judgment signal z output terminal 704. Also shown in FIG. 105 are a
counting value c of the chip counter 70, a non-zero code w obtained by the
data converter 71, a spread code pp, a result mo of the multiplier 73 and
a user identification No. i. The hopping code y and the binary judgment
signal z respectively supplied through the terminals 703, 704 are supplied
to a frequency synthesizer having a choice of at least 16 carrier
frequencies and provided with a carrier non-transmission mode.
FIG. 106 shows the operation of the FH code judgment unit 75. In this
example, Q is equal to 17 and the value of the hopping code y obtained by
the adder 74 may be in the range from 0 to 16. For the hopping code y, the
FH code judgment unit 75 judges only predetermined M (=16=2.sup.4)
different values, i.e., y=0, 1, . . . , 15, as effective (z=1), and judges
other values as ineffective (z=0). The frequency synthesizer connected to
the FH code generator 700 in FIG. 105, is brought to the carrier
non-transmission mode when z is equal to 0, and transmits the carrier
frequency corresponding to the hopping code y when z is equal to 1.
According to the arrangement in FIG. 105, there is positively utilized the
feature of the FH mode that decoding can be made even though some of a
plurality of carrier frequencies are lost. That is, with a portion of
hopping codes y invalidated, there are used hopping codes in number of
two's power, thus preventing the frequency synthesizer from being
complicated in hardware.
In each of the examples of the FH code generator, the number of values that
a data can present, is set to M=2k (k is a positive integer) and the
number Q of the elements of a Galois field is set to p.sup.r (>M) in which
p is a prime number and r is a positive integer. The number of chips L is
an integer not less than 2 and not greater than p.sup.r -1. Preferably,
the number Q of the elements of a Galois field is set to the minimum value
out of all the values Q each of which satisfies the condition of Q=p.sup.r
(>M).
FIG. 107 shows examples of M and Q which satisfy the condition
above-mentioned. In the Q column in FIG. 107, for example, "5" is a prime
number 5 to the first power, "9" is a prime number 3 to the second power,
and "17" is a prime number 17 to the first power. According to FIG. 107,
for example, when M is equal to 512, Q is equal to 521. Accordingly, the
number of invalidated hopping codes is as small as 9. More specifically,
by maximizing the probability of hopping codes being validated, the
maximum system reliability can be achieved. That is, decoding is
practically sufficiently made even though the quality is somewhat
deteriorated as compared with the case where all Q hopping codes are used.
* * * * *
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