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| United States Patent |
6,535,715
|
|
Dapper
, et al.
|
March 18, 2003
|
Hybrid/fiber coax video and telephony communication system with poly-phase
filtering
Abstract
A communication system is described. The communication system includes a
distribution network between a head end terminal and at least one remote
unit. The head end terminal receives upstream telephony and control data
modulated on a plurality of orthogonal carriers in a frequency bandwidth
over the distribution network. The head end terminal includes a polyphase
filter that filters at least the upstream telephony data to provide
ingress protection for the modulated orthogonal carriers. The
communication further includes a service unit multicarrier modem
associated with the at least one remote unit and operatively connected to
the distribution network for modulating at least upstream telephony
information on at least one carrier orthogonal at the head end terminal to
at least one other carrier in the frequency bandwidth.
| Inventors:
|
Dapper; Mark J. (Cincinnati, OH);
Geile; Michael J. (Batavia, OH)
|
| Assignee:
|
ADC Telecommunications, Inc. (Eden Prairie, MN)
|
| Appl. No.:
|
396308 |
| Filed:
|
September 15, 1999 |
| Current U.S. Class: |
455/3.05; 370/210; 370/484; 455/3.01 |
| Intern'l Class: |
H04J 011/00; H04J 001/00; H04H 001/00 |
| Field of Search: |
455/5.1
370/210,484
|
References Cited [Referenced By]
U.S. Patent Documents
| 4977593 | Dec., 1990 | Ballance.
| |
| 5027426 | Jun., 1991 | Choicca, Jr.
| |
| 5153763 | Oct., 1992 | Pidgeon.
| |
| 5195092 | Mar., 1993 | Wilson et al.
| |
| 5262883 | Nov., 1993 | Pidgeon.
| |
| 5299192 | Mar., 1994 | Guo et al.
| |
| 5303229 | Apr., 1994 | Withers et al.
| |
| 5323391 | Jun., 1994 | Harrison | 370/210.
|
| 5499047 | Mar., 1996 | Terry et al.
| |
| 5499241 | Mar., 1996 | Thompson et al.
| |
| 5512937 | Apr., 1996 | Beierle.
| |
| 5553064 | Sep., 1996 | Paff et al.
| |
| 5610944 | Mar., 1997 | Mau et al. | 370/484.
|
| 5745837 | Apr., 1998 | Fuhrmann | 455/5.
|
| 5995539 | Nov., 1999 | Miller.
| |
| 6091932 | Jul., 2000 | Langlais | 455/5.
|
| Foreign Patent Documents |
| 0 499 065 | Aug., 1992 | EP.
| |
| 595710 | May., 1994 | EP.
| |
| 0 613 266 | Aug., 1994 | EP.
| |
| 0 695 054 | Jan., 1996 | EP.
| |
| 01-319319 | Dec., 1989 | JP.
| |
| WO 96/10303 | Apr., 1996 | WO.
| |
| WO 96/24989 | Aug., 1996 | WO.
| |
Other References
Sierens, Mestdagh, Van Der Plas, Vandewege, Depovere, Debie, "Subcarrier
multiple access for passive optical networks and comparison to other
mutable access techniques", GLOBECOM 91, Feb. 12, 1991, IEEE.*
Stark, "AM Transmission on Fiber" Communications Engineering and Design.
Apr. 1988.*
ANT Communications Engineering, "Communications Engineering reports," May
1984.*
Chiddix, J.A., et al., "Fiber Backbone: A Proposal for an Evolutionary CATV
Network Architecture," 1988 NCTA Technical Papers, 73-82, (1988).
Engstrom, B., et al., "A System for Test of Multiaccess Methods based on
OFDM," Proceedings of the 1994 IEEE 44th Vehicular Technology Conference,
vol. 3, Stockholm, Sweden, 1843-1845, (Jun. 8-10, 1994).
Fliege, N.J., "Orthogonal Multiple Carrier Data Transmission," European
Transactions on Telecommunications and Related Technologies, 3, 255-264,
(May-Jun., 1992).
Wintermantel, J., et al., "A Class of DFT Polyphase Filter Banks for
Orthogonal Multliple Carrier Data Transmission," Proceedings of the 1992
IEEE International Symposium on Circuits and Systems, vol. 3, San Diego,
CA, 1332-1335, (May 10-13, 1992).
|
Primary Examiner: Cumming; William
Attorney, Agent or Firm: Fogg and Associates, LLC
Parent Case Text
CROSS REFERENCE TO RELATED CASES
This Application is a divisional of U.S. application Ser. No. 08/673,002
filed Jun. 28, 1996 (now U.S. Pat. No. 6,334,219) which is a
continuation-in-part of U.S. application Ser. Nos. 08/650,408 filed May
20, 1996 (abandoned), 08/457,295 filed Jun. 1, 1995 (abandoned),
08/384,659 filed Feb. 6, 1995 (abandoned), and 08/457,317 filed Jun. 1,
1995 (abandoned), which applications are incorporated by reference. This
application is related to U.S. application Ser. Nos. 08/311,964 filed Sep.
26, 1994 (abandoned), 08/455,340 filed May 31, 1995 (abandoned),
08/455,059 filed May 31, 1995 (abandoned), 08/457,294 filed Jun. 1, 1995
(abandoned), 08/457,110 filed Jun. 1, 1995 (abandoned), 08/456,871 filed
Jun. 1, 1995 (abandoned), 08/457,022 filed Jun. 1, 1995 (abandoned), and
08/457,037 filed Jun. 1, 1995 (abandoned), which applications are
incorporated by reference.
Claims
What is claimed is:
1. A communication system, comprising:
a distribution network between a head end terminal and at least one remote
unit, the head end terminal for receipt of upstream telephony information
and upstream control data in a frequency bandwidth over the distribution
network;
at least one service unit multicarrier means, each associated with at least
one remote unit and operatively connected to the distribution network for
modulating at least upstream telephony information on at least one carrier
orthogonal at the head end terminal to at least one other carrier in the
frequency bandwidth;
service unit controller means operatively connected to the service unit
multicarrier means for controlling the modulation performed by the service
unit multicarrier means; and
wherein the head end terminal includes:
head end multicarrier means for demodulating at least upstream telephony
information modulated on a plurality of orthogonal carriers in the
frequency bandwidth, the head end multicarrier means including polyphase
filter means for filtering the at least upstream telephony information
modulated on the plurality of orthogonal carriers to provide ingress
protection for the modulated orthogonal carriers; and
head end controller means operatively connected to the head end
multicarrier means for controlling receipt of the upstream control data
and upstream telephony information.
2. The system according to claim 1, wherein the plurality of orthogonal
carriers in the frequency bandwidth include a plurality of telephony
information channels for transmission of upstream telephony information
after modulation of upstream telephony information thereon and at least
one control channel associated with the plurality of telephony information
channels for transmission of upstream control data after modulation of
upstream control data thereon.
3. The system according to claim 1, wherein
the head end terminal further includes downstream multicarrier means for
modulating at least downstream control information on a carrier in an
additional frequency bandwidth over the distribution network,
the head end controller means is further operatively connected to the
downstream multicarrier means for controlling transmission of the at least
downstream control data and for generating a command for transmission by
the head end terminal as downstream control information for the remote
unit to utilize at least one different carrier for upstream transmission
in the frequency bandwidth,
the at least one service unit multicarrier means is further operatively
connected to the distribution network for demodulating the command
transmitted as downstream control information modulated on the at least
one carrier in the additional frequency bandwidth, and
the service unit controller means operatively connected to the service unit
multicarrier means for controlling the demodulation performed by the
service unit multicarrier means and for receiving the command and
controlling the at least one service unit to transmit utilizing the
different carrier as instructed by the head end controller.
4. A method of polyphase filtering in a communication system, comprising
the steps of:
receiving a plurality of orthogonal carriers having modulated telephony
information thereon, the plurality of orthogonal carriers including a
first and second plurality of noncontiguous channel sets;
filtering the first plurality of noncontiguous channel sets and passing a
first plurality of channels of each channel set of the first plurality of
noncontiguous channel sets; and
filtering the second plurality of noncontiguous channel sets and passing a
second plurality of channels;
wherein passing the second plurality of channels includes passing channels
of each channel set of the second plurality of noncontiguous channel sets
and passing channels not passed when passing the filtered channels of the
first plurality of non- contiguous channel sets.
5. A communication system, comprising:
a distribution network between a head end terminal and at least one remote
unit, the head end terminal receives upstream telephony and control data
modulated on a plurality of orthogonal carriers in a frequency bandwidth
over the distribution network, wherein the head end terminal includes a
polyphase filter that filters at least the upstream telephony data to
provide ingress protection for the modulated orthogonal carriers; and
a service unit multicarrier modem associated with the at least one remote
unit and operatively connected to the distribution network for modulating
at least upstream telephony information on at least one carrier orthogonal
at the head end terminal to at least one other carrier in the frequency
bandwidth.
6. The system of claim 5, wherein the plurality of orthogonal carriers in
the frequency bandwidth include a plurality of telephony information
channels for transmission of upstream telephony information after
modulation of upstream telephony information thereon and at least one
control channel associated with the plurality of telephony information
channels for transmission of upstream control data after modulation of
upstream control data thereon.
7. The system of claim 5, wherein the head end terminal further includes:
a head end multicarrier modem for modulating at least downstream control
information on a carrier in an additional frequency bandwidth over the
distribution network; and
a head end controller operatively connected to the head end multicarrier
modem for controlling transmission of the at least downstream control data
and for generating a command for transmission by the head end terminal as
downstream control information for the remote unit to utilize at least one
different carrier for upstream transmission in the frequency bandwidth.
8. The system of claim 7, wherein the service unit multicarrier modem is
further operatively connected to the distribution network for demodulating
the command transmitted as downstream control information modulated on the
at least one carrier in the additional frequency bandwidth.
9. The system of claim 8, further comprising a service unit controller
operatively connected to the service unit multicarrier modem for
controlling the demodulation performed by the service unit multicarrier
modem and for receiving the command and controlling the at least one
remote unit to transmit utilizing the different carrier as instructed by
the head end controller.
10. The system of claim 5, wherein the polyphase filter includes a first
and second polyphase filter.
11. The communication system of claim 10, wherein the first polyphase
filter filters a first plurality of channel sets and passes a first
plurality of at least telephony channels within each channel set of the
first plurality of channel sets and wherein the second polyphase filter
filters a second plurality of channel sets and passes a second plurality
of at least telephony channels within each channel set of the second
plurality of channel sets, the first and second polyphase filter offset
from one another such that all at least telephony channels of the first
and second plurality of channels sets are passed.
12. The communication system of claim 5, wherein the polyphase filter
includes at least two staggered polyphase filters.
13. A communication system, comprising:
a head end terminal;
at least one remote unit;
a distribution network, coupled between the head end terminal and the at
least one remote unit;
the head end terminal including:
a head end multicarrier modem adapted to demodulate signals modulated on a
plurality of orthogonal carriers in a first frequency bandwidth,
a head end controller operatively connected to the head end multicarrier
modem for controlling receipt of signals; and
wherein the head end multicarrier modem includes a polyphase filter adapted
to selectively filter the received signals to provide ingress protection
for the modulated orthogonal carriers;
the at least one remote unit including:
a second multicarrier modem, operatively connected to the distribution
network for modulating signals on at least one carrier orthogonal at the
head end terminal to at least one other carrier in the frequency
bandwidth; and
a second controller operatively connected to the second multicarrier modem
for controlling the modulation performed by the second multicarrier modem.
14. The communication system of claim 13, wherein the polyphase filter
includes a first and second polyphase filter.
15. The communication system of claim 14, wherein the first polyphase
filter filters a first plurality of channel sets and passes a first
plurality of at least telephony channels within each channel set of the
first plurality of channel sets and wherein the second polyphase filter
filters a second plurality of channel sets and passes a second plurality
of at least telephony channels within each channel set of the second
plurality of channel sets, the first and second polyphase filter offset
from one another such that all at least telephony channels of the first
and second plurality of channels sets are passed.
16. The communication system of claim 13, wherein the polyphase filter
includes at least two overlapping polyphase filters.
Description
FIELD OF THE INVENTION
The present invention relates generally to the field of communication
systems. More particularly, the present invention relates to communication
systems with multicarrier telephony transport
BACKGROUND OF THE INVENTION
Two information services found in households and businesses today include
television, or video, services and telephone services. Another information
service involves digital data transfer which is most frequently
accomplished using a modem connected to a telephone service. All further
references to telephony herein shall include both telephone services and
digital data transfer services.
Characteristics of telephony and video signals are different and therefore
telephony and video networks are designed differently as well. For
example, telephony information occupies a relatively narrow band when
compared to the bandwidth for video signals. In addition, telephony
signals are low frequency whereas NTSC standard video signals are
transmitted at carrier frequencies greater than 50 MHz. Accordingly,
telephone transmission networks are relatively narrow band systems which
operate at audio frequencies and which typically serve the customer by
twisted wire drops from a curb-side junction box. On the other hand, cable
television services are broad band and incorporate various frequency
carrier mixing methods to achieve signals compatible with conventional
very high frequency television receivers. Cable television systems or
video services are typically provided by cable television companies
through a shielded cable service connection to each individual home or
business.
One attempt to combine telephony and video services into a single network
is described in U.S. Pat. No. 4,977,593 to Balance entitled "Optical
Communications Network." Balance describes a passive optical
communications network with an optical source located in a central
station. The optical source transmits time division multiplexed optical
signals along an optical fiber and which signals are later split by a
series of splitters between several individual fibers servicing
outstations. The network allows for digital speech data to be transmitted
from the outstations to the central station via the same optical path. In
addition, Balance indicates that additional wavelengths could be utilized
to add services, such as cable television, via digital multiplex to the
network.
A 1988 NCTA technical paper, entitled "Fiber Backbone: A Proposal For an
Evolutionary Cable TV network Architecture," by James A. Chiddix and David
M. Pangrac, describes a hybrid optical fiber/coaxial cable television
(CATV) system architecture. The architecture builds upon existing coaxial
CATV networks. The architecture includes the use of a direct optical fiber
path from a head end to a number of feed points in an already existing
CATV distribution system.
U.S. Pat. No. 5,153,763 to Pidgeon, entitled "CATV Distribution Networks
Using Light Wave Transmission Lines," describes a CATV network for
distribution of broad band, multichannel CATV signals from a head end to a
plurality of subscribers. Electrical to optical transmitters at the head
end and optical to electrical receivers at a fiber node launch and receive
optical signals corresponding to broad band CATV electrical signals.
Distribution from the fiber node is obtained by transmitting electrical
signals along coaxial cable transmission lines. The system reduces
distortion of the transmitted broad band CATV signals by block conversion
of all or part of the broad band of CATV signals to a frequency range
which is less than an octave. Related U.S. Pat. No. 5,262,883 to Pidgeon,
entitled "CATV Distribution Networks Using Light Wave Transmission Lines,"
further describes the distortion reducing system.
Although the above-mentioned networks describe various concepts for
transmitting broad band video signals over various architectures, which
may include hybrid optical fiber/coax architectures, none of these
references describe a cost effective, flexible, communications system for
telephony communications. Several problems are inherent in such a
communication system.
One such problem is the need to optimize the bandwidth used for
transporting data so that the bandwidth used does not exceed the allotted
bandwidth. Bandwidth requirements are particularly critical in multi-point
to point communication where multiple transmitters at remote units must be
accommodated such that allotted bandwidth is not exceeded.
A second problem involves power consumption of the system. The
communication system should minimize the power used at the remote units
for the transport of data, as the equipment utilized at the remote units
for transmission and reception may be supplied by power distributed over
the transmission medium of the system.
Another problem arises from a fault in the system preventing communication
between a head end and multiple remote units of a multi-point to point
system. For example, a cut transmission line from a head end to many
remote units may leave many users without service. After the fault is
corrected, it is important bring as many remote units back into service as
quickly as possible.
Data integrity must also be addressed. Both internal and external
interference can degrade the communication. Internal interference exists
between data signals being transported over the system. That is,
transported data signals over a common communication link may experience
interference therebetween, decreasing the integrity of the data Ingress
from external sources can also effect the integrity of data transmissions.
A telephony communication network is susceptible to "noise" generated by
external sources, such as HAM radio. Because such noise can be
intermittent and vary in intensity, a method of transporting data over the
system should correct or avoid the presence of such ingress.
These problems and others as will become apparent from the description to
follow, present a need for an enhanced communication system. Moreover,
once the enhanced system is described, a number of practical problems in
its physical realization are presented and overcome.
Another embodiment provides a method and apparatus for a fast Fourier
transform. This invention relates to the field of electronic communication
systems, and more specifically to an improved method and apparatus for
providing a fast Fourier transform ("FFT").
There are many advanced digital signal-processing applications requiring
analysis of large quantities of data in short time periods, especially
where there is interest in providing "real time" results. Such
applications include signal processing in modems which use OFDM
(orthogonal frequency division multiplexing). In order to be useful in
these and other applications, Discrete Fourier Transform (DFT) or Fast
Fourier Transform (FFT) signal processors must accommodate large numbers
of transforms, or amounts of data, in very short processing times, often
called high data throughput.
In addition to the speed and data-throughput requirements, power
consumption is a major concern for many applications. In some
signal-processing applications, power is supplied by portable generation
or storage equipment, such as batteries, where the ultimate power
available is limited by many environment. In such applications, processor
power consumption must be as low as possible. One useful measure of
utility or merit for FFT processors is the energy dissipation per
transform point. Ultimately, one key problem with any FFT processor is the
amount of power consumed per transform. Generally, high-performance,
efficient FFT processors exhibit energy dissipations per transform in the
range of 100 to 1000 times log.sub.2 N nanojoules, where N is the number
of points in a given transform. As a consequence, reasonably large
transforms required to process large arrays of data, result in large power
consumption.
Machine-implemented computation of an FFT is often simplified by cascading
together a series of simple multiply-and-add stages. When a recursive
process is used, data circulates through a single stage and the
computational structure of the stage is made variable for each
circulation. Each circulation through the stage is referred to as a
"pass".
A plurality of computational elements, each known as a radix-r butterfly,
may be assembled to define a single stage for carrying out a particular
pass. A radix-r butterfly receives r input signals and produces a
corresponding number of r output signals, where each output signal is the
weighted sum of the r input signals. The radix number, r, in essence,
defines the number of input components which contribute to each output
component
By way of example, a radix-2 butterfly receives two input signals and
produces two output signals. Each output signal is the weighted sum of the
two input signals. A radix-4 butterfly receives four input signals and
produces four corresponding output signals. Each output signal of the
radix-4 butterfly constitutes a weighted sum of the four input signals.
Completion of an N-point Fast Fourier Transform (FFT) requires that the
product of the butterfly radix values, taken over the total number of
stages or passes, equals the total point count, N. Thus, a 64-point FFT
can be performed by one radix-64 butterfly, or three cascaded stages where
each stage has sixteen radix-4 butterflies (the product of the radix
values for stage-1 and stage-2 and stage-3 is 4.times.4.times.4=64), or
six cascaded stages where each of the six stages comprises 32 radix-2
butterflies (the product of the radix values for stage-1 through stage-6
is 2.times.2.times.2.times.2.times.2.times.2=64).
A multi-stage or multi-pass FFT process can be correctly carried out under
conditions where the number of butterfly elements changes from one pass
(or stage) to the next and the radix value, r, of the butterfly elements
also changes from one pass (or stage) to the next. A paper by Gordon
DeMuth, "ALGORITHMS FOR DEFINING MIXED RADIX FFT FLOW GRAPHS", IEEE
Transactions on Acoustics, Speech, and Signal Processing, Vol 37, No. 9,
Sep. 1989, Pages 1349-1358, describes a generalized method for performing
an FFT with a mixed-radix system. A mixed-radix system is one where the
radix value, r, in one stage or pass is different from that of at least
one other stage or pass.
An advantage of a mixed-radix computing system is that it can be "tuned" to
optimize the signal-to-noise ratio of the transform (or more correctly
speaking, to minimize the accumulated round-off error of the total
transform) for each particular set of circumstances. By way of example, it
is advantageous in one environment to perform a 512-point FFT using the
mixed-radix sequence: 4,4,4,4,2. In a different environment, it may be
more advantageous to use the mixed-radix sequence: 4, 2, 4, 4, 4.
Round-off error varies within a machine of finite precision as a function
of radix value and the peak signal magnitudes that develop in each stage
or pass.
In addition, it may be advantageous to scale intermediate results between
each stage or pass, in order to minimize round-off errors and the problem
of overflow. Further, it may be advantageous to vary the amount of scaling
performed between each pass, e.g., either to scale by 1/4 between each
radix-4 stage or to scale by 1/2 for some stages and 1/8 for other stages.
Heretofore, FFT processors generally fetched data values from their working
storage in a serial manner, thus limiting the speed which could be
obtained. Further, current FFT processors generally were limited in speed
by loading the working storage with input values, then processing the data
in the working storage, then unloading the result values.
There are many advanced digital signal-processing applications requiring
analysis of large quantities of data in short time periods, especially
where there is interest in providing "real time" results. Such
applications include signal processing in modems which use OFDM
(orthogonal frequency division multiplexing).
One need in the art is for an accurate analog-to-digital conversion (ADC)
at moderate frequencies having limited bandwidth. One technology known in
the art is the "Sigma-Delta" ADC which provides very good resolution (high
number of bits in the digital result), but only for signals whose
converted signal bandwidth is low.
Another need is for an ADC which provides bandwidth-limited digital I and Q
signals (representing amplitude and quadrature) for a 200 kHz bandwidth
received analog modem signal, wherein the digital result has very high
resolution and accuracy.
What is needed is a method and apparatus which addresses the above problems
in the art.
SUMMARY OF THE INVENTION
Embodiments of the present invention provide a communication system. In one
embodiment, the communication system includes a distribution network
between a head end terminal and at least one remote unit. The head end
terminal receives upstream telephony and control data modulated on a
plurality of orthogonal carriers in a frequency bandwidth over the
distribution network. The head end terminal includes a polyphase filter
that filters at least the upstream telephony data to provide ingress
protection for the modulated orthogonal carriers. The communication
further includes a service unit multicarrier modem associated with the at
least one remote unit and operatively connected to the distribution
network for modulating at least upstream telephony information on at least
one carrier orthogonal at the head end terminal to at least one other
carrier in the frequency bandwidth.
DESCRIPTION OF THE DRAWINGS
FIG. 1 shows a block diagram of a communication system in accordance with
the present invention utilizing a hybrid fiber/coax distribution network;
FIG. 2 is an alternate embodiment of the system of FIG. 1;
FIG. 3 is a detailed block diagram of a host digital terminal (HDT) with
associated transmitters and receivers of the system of FIG. 1;
FIG. 4 is a block diagram of the associated transmitters and receivers of
FIG. 3;
FIG. 5 is a block diagram of an optical distribution node of the system of
FIG. 1;
FIGS. 6, 7 are embodiments of frequency shifters for use in the optical
distribution node of FIG. 5 and the telephony upstream receiver of FIG. 4,
respectively;
FIG. 8 is a general block diagram of an integrated service unit (ISU) such
as a home integrated service unit (HISU) or a multiple integrated service
unit (MISU) of FIG. 1;
FIGS. 9, 10, 11 show data frame structures and frame signaling utilized in
the HDT of FIG. 3;
FIG. 12 is a general block diagram of a coax master card (CXMC) of a coax
master unit (CXMU) of FIG. 3;
FIG. 13 shows a spectral allocation for a first transport embodiment for
telephony transport in the system of FIG. 1;
FIG. 14 shows a mapping diagram for QAM modulation;
FIG. 15 shows a mapping diagram for BPSK modulation;
FIG. 16 shows a subband diagram for the spectral allocation of FIG. 13;
FIGS. 17, 18 show alternative mapping diagrams or constellations for QAM
modulation;
FIG. 19 shows a timing diagram of an identification and synchronization
process;
FIG. 20 shows a timing diagram of a burst identification and
synchronization process;
FIG. 21 is a block diagram of a master coax card (MCC) downstream
transmission architecture of the CXMU for the first transport embodiment
of the system of FIG. 1;
FIG. 22 is a block diagram of a coax transport unit (CXTU) downstream
receiver architecture of an MISU for the first transport embodiment of the
system of FIG. 1;
FIG. 23 is a block diagram of a coax home module (CXHM) downstream receiver
architecture of an HISU for the first transport embodiment of the of the
system of FIG. 1;
FIG. 24 is a block diagram of a CXHM upstream transmission architecture
associated with the CXHM downstream receiver architecture of FIG. 23;
FIG. 25 is a block diagram of a CXTU upstream transmission architecture
associated with the CXTU downstream receiver architecture of FIG. 22;
FIG. 26 is a block diagram of an MCC upstream receiver architecture
associated with the MCC downstream transmission architecture of FIG. 21;
FIG. 27 is a flow diagram of a acquisition distributed loop routine for use
with the system of FIG. 1;
FIG. 28 is a flow diagram of a tracking distributed loop architecture
routine for use with the system of FIG. 1;
FIG. 29 shows a magnitude response of a polyphase filter bank of the MCC
upstream receiver architecture of FIG. 26;
FIG. 30 is an enlarged view of part of the magnitude response of FIG. 29;
FIG. 31 is a block diagram of an ingress filter structure and FFT of the
MCC upstream receiver architecture of FIG. 26;
FIG. 32 is a block diagram of a polyphase filter structure of the ingress
filter structure and FFT of FIG. 31;
FIG. 33 is a block diagram of a carrier, amplitude, timing recovery block
of the downstream receiver architectures of the first transport
embodiment;
FIG. 34 is a block diagram of a carrier, amplitude, timing recovery block
of the MCC upstream receiver architecture of the first transport
embodiment;
FIG. 35 is a block diagram of internal equalizer operation for the receiver
architectures of the first transport embodiment;
FIG. 36 is a spectral allocation of a second transport embodiment for
transport in the system of FIG. 1;
FIG. 37 is a block diagram of an MCC modem architecture of the CXMU for the
second transport embodiment of the system of FIG. 1;
FIG. 38 is a block diagram of a subscriber modem architecture of the HISU
for the second transport embodiment of the system of FIG. 1;
FIG. 39 is a block diagram of a modem of the subscriber modem architecture
of FIG. 38;
FIG. 40 is a block diagram for channel monitoring used in the system of
FIG. 1;
FIGS. 41, 42, 43 are flow diagrams for error monitor portions of channel
monitor routines of FIG. 40;
FIG. 44 is an alternate flow diagram for the diagram of FIG. 42;
FIG. 45 is a flow diagram for a background monitor portion of the channel
monitor routines of FIG. 40;
FIG. 46 is a flow diagram for a backup portion of the channel monitor
routines of FIG. 40;
FIGS. 47, 48 are a flow diagram of an acquisition distributed loop routine
for use with another embodiment of the system of FIG. 1;
FIG. 49 is a flow diagram of a downstream tracking loop for use with the
embodiment of FIGS. 47 and 48.
FIG. 50 is a flow diagram of an upstream tracking loop for use with the
embodiment of FIGS. 47 and 48.
FIG. 51 is a block diagram showing the locking of all clocks within a
system.
FIGS. 52, 53 depict phase diagrams of symbol waveforms in an embodiment of
the invention.
FIGS. 54, 55, 56, 57 describe error rates and message-encoding methods for
use in a system according to the invention.
FIG. 58 is a block diagram of a scrambler for use in the invention.
FIG. 59 is a block diagram of a control circuit for a CXMU of an HDT in a
telecommunications system;
FIGS. 60, 61, 62 are flow charts that illustrate methods for assigning
subbands and allocating payload channels in a telecommunications system
that uses a multi-carrier communication scheme;
FIGS. 63, 64, 65, 66, 67 are frequency spectrum diagrams that illustrate
examples of assigning service units to subbands;
FIG. 68 is a flow chart that illustrates error monitoring by the channel
manager;
FIG. 69 is a flow chart that illustrates a method for allocating an ISU
data-link (IDL) channel in a telecommunications system;
FIG. 70 is a block diagram of FFT system 2100;
FIG. 71 is a block diagram of modem 2400 which includes a FFT system 2100
configured to perform an IFFT in transmitter section 2401 and another FFT
system 2100 configured to perform an FFT in receiver section 2402;
FIG. 72 is a block diagram of three logical banks of RAM: an input RAM
2251, an output RAM 2253, and a conversion RAM 2252;
FIG. 73 is a block diagram of one embodiment of a physical implementation
which provides the function of input RAM 2241, conversion RAM 2242, and
output RAM 2243;
FIG. 74 is a block diagram of one embodiment of a dual radix core 2600;
FIGS. 75, 76, 77, 78, 79, 80, 81, 82 together form a table showing the
order of calculations for a "normal butterfly sub-operation";
FIGS. 83, 84, 85, 86, 87, 88, 89, 90 together form a table showing the
order of calculations for a "transposed butterfly sub-operation";
FIG. 91 is a block diagram of one embodiment of dual-radix core 2600
showing the nomenclature used for the products output by multipliers 2620
through 2627 and for adder-subtractor-accumulators 2633;
FIG. 92 is a block diagram of one embodiment of an
adder-subtractor-accumulator 2633;
FIG. 93 is a block diagram of modem 2400 which includes a Sigma-Delta ADC
and decimator system to drive FFT system 2100;
FIG. 94 is a more detailed block diagram of modem receiver 2402;
FIG. 95 is a detailed block diagram of one embodiment of a Sigma-Delta
converter 2840;
FIG. 96 is an overall schematic diagram of the data delivery transport
system according to the present invention;
FIG. 97 is a simplified block diagram of the head-end terminal 12 of the
system 500 according to the present invention;
FIG. 98 illustrates a Personal Cable Data Modem (PCDM) 540 and a Data Modem
Service Module (DMSM) 550;
FIG. 99 illustrates in greater detail a PCDM 540;
FIG. 100 illustrates a Data Modem Channel Unit (DMCU) 560;
FIG. 101 shows a graph of average bandwidth per user as a function of the
number of users for the system 500 according to the present invention;
FIG. 102 is a simplified block diagram of the data transport and framing of
the system 500 according to the present invention;
FIG. 103 illustrates a Local Area Network Unit (LANU) 580 according to the
present invention;
FIG. 104 illustrates in more detail a DMSM 550 according to the present
invention;
FIG. 105 illustrates in more detail a DMCU 560 according to the present
invention;
FIGS. 106, 107, 108, 109 illustrate the call setup for a data connection on
the system 500 according to the present invention;
FIG. 110 illustrates a call termination sequence on the system 500
according to the present invention;
FIG. 111 illustrates the software of a LANU 580 according to the present
invention;
FIG. 112 illustrates a PCDM 620 adapted for asymmetrical data delivery;
FIG. 113 illustrates the head-end configuration for asymmetrical data
delivery according to the present invention;
FIGS. 114, 115 illustrate another alternate embodiment of the invention
wherein digital video is received over an ATM network and transmitted over
a modified form of system 10/500;
FIG. 116 shows a block diagram of a hybrid fiber/coax network in accordance
with the present invention;
FIG. 117 is a block diagram of a head end host distribution terminal of the
network of FIG. 116;
FIG. 118 is a block diagram of an optical distribution node of the network
of FIG. 116;
FIG. 119 is a block diagram of a home coaxial line unit of the network of
FIG. 116;
FIG. 120 is a block diagram of an alternative embodiment for transmission
from the head end to the optical distribution nodes in accordance with the
present invention;
FIG. 121 is a block diagram of an impulse shaping technique utilized in
accordance with the present invention;
FIG. 122 is a block diagram of an alternative embodiment of the optical to
electrical converter of the head end host distribution terminal of FIG.
117;
FIG. 123 is a block diagram of an alternative embodiment of the head end
host distribution terminal of FIG. 117.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
The communication system 10, as shown in FIG. 1, of the present invention
is an access platform primarily designed to deliver residential and
business telecommunication services over a hybrid fiber-coaxial (HFC)
distribution network 11. The system 10 is a cost-effective platform for
delivery of telephony and video services. Telephony services may include
standard telephony, computer data and/or telemetry. In addition, the
present system is a flexible platform for accommodating existing and
emerging services for residential subscribers.
The hybrid fiber-coaxial distribution network 11 utilizes optical fiber
feeder lines to deliver telephony and video service to a distribution node
18 (referred to hereinafter as the optical distribution node (ODN))
remotely located from a central office or a head end 32. From the ODNs 18,
service is distributed to subscribers via a coaxial network. Several
advantages exist by utilizing the HFC-based communication system 10. By
utilizing fiber installed in the feeder, the system 10 spreads the cost of
optoelectronics across hundreds of subscribers. Instead of having a
separate copper loop which runs from a distribution point to each
subscriber ("star" distribution approach), the system 10 implements a
bused approach where a distribution coaxial leg 30 passes each home and
subscribers "tap" the distribution coaxial leg 30 for service. The system
10 also allows non-video services to be modulated for transmission using
more cost-effective RF modem devices in dedicated portions of the RF
spectrum. Finally, the system 10 allows video services to be carried on
existing coaxial facilities with no additional subscriber equipment
because the coaxial distribution links can directly drive existing
cable-ready television sets.
It should be apparent to one skilled in the art that the modem transport
architecture described herein and the functionality of the architecture
and operations surrounding such architecture could be utilized with
distribution networks other than hybrid fiber coax networks. For example,
the functionality may be performed with respect to wireless systems.
Therefore, the present invention contemplates use of such systems in
accordance with the accompanying claims.
The system 10 includes host digital terminals 12 (HDTs) which implement all
common equipment functions for telephony transport, such as network
interface, synchronization, DS0 grooming, and operations, administration,
maintenance and provisioning (OAM&P) interfaces, and which include the
interface between the switching network and a transport system which
carries information to and from customer interface equipment such as
integrated service units 100 (ISUs). Integrated services units (ISUs) 100,
such as home integrated service units (HISUs) 68 or multiple user
integrated service units (MISUs) 66, which may include a business
integrated service unit as opposed to a multiple dwelling integrated
service unit, implement all customer interface functions and interface to
the transport system which carries information to and from the switched
network. In the present system, the HDT 12 is normally located in a
central office and the ISUs 100 are remotely located in the field and
distributed in various locations. The HDT 12 and ISUs 100 are connected
via the hybrid fiber-coax distribution network 11 in a multi-point to
point configuration. In the present system, the modem functionality
required to transport information over the HFC distribution network 11 is
performed by interface equipment in both the HDT 12 and the ISUs 100. Such
modem functionality is performed utilizing orthogonal frequency division
multiplexing.
The communication system shall now be generally described with reference to
FIGS. 1, 3 and 8. The primary components of system 10 are host digital
terminals (HDTs) 12, video host distribution terminal (VHDT) 34, telephony
downstream transmitter 14, telephony upstream receiver 16, the hybrid
fiber coax (HFC) distribution network 11 including optical distribution
node 18, and integrated service units 66, 68 (shown generally as ISU 100
in FIG. 8) associated with remote units 46. The HDT 12 provides telephony
interface between the switching network (noted generally by trunk line 20)
and the modem interface to the HFC distribution network for transport of
telephony information. The telephony downstream transmitter 14 performs
electrical to optical conversion of coaxial RF downstream telephony
information outputs 22 of an HDT 12, shown in FIG. 3, and transmits onto
redundant downstream optical feeder lines 24. The telephony upstream
receiver 16 performs optical to electrical conversion of optical signals
on redundant upstream optical feeder lines 26 and applies electrical
signals on coaxial RF upstream telephony information inputs 28 of HDT 12.
The optical distribution node (ODN) 18 provides interface between the
optical feeder lines 24 and 26 and coaxial distribution legs 30. The ODN
18 combines downstream video and telephony onto coaxial distribution legs
30. The integrated services units provide modem interface to the coaxial
distribution network and service interface to customers.
The HDT 12 and ISUs 100 implement the telephony transport system
modulator-demodulator (modem) functionality. The HDT 12 includes at least
one RF MCC modem 82, shown in FIG. 3 and each ISU 100 includes an RF ISU
modem 101, shown in FIG. 8. The MCC modems 82 and ISU modems 101 use a
multi-carrier RF transmission technique to transport telephony
information, such as DS0+ channels, between the HDT 12 and ISUs 100. This
multi-carrier technique is based on orthogonal frequency division
multiplexing (OFDM) where a bandwidth of the system is divided up into
multiple carriers, each of which may represent an information channel.
Multi-carrier modulation can be viewed as a technique which takes
time-division multiplexed information data and transforms it to
frequency-division multiplexed data The generation and modulation of data
on multiple carriers is accomplished digitally, using an orthogonal
transformation on each data channel. The receiver performs the inverse
transformation on segments of the sampled waveform to demodulate the data
The multiple carriers overlap spectrally. However, as a consequence of the
orthogonality of the transformation, the data in each carrier can be
demodulated with negligible interference from the other carriers, thus
reducing interference between data signals transported. Multi-carrier
transmission obtains efficient utilization of the transmission bandwidth,
particularly necessary in the upstream communication of a multi-point to
point system. Multi-carrier modulation also provides an efficient means to
access multiple multiplexed data streams and allows any portion of the
band to be accessed to extract such multiplexed information, provides
superior noise immunity to impulse noise as a consequence of having
relatively long symbol times, and also provides an effective means for
eliminating narrowband interference by identifying carriers which are
degraded and inhibiting the use of these carriers for data transmission
(such channel monitoring and protection is described in detail below).
Essentially, the telephony transport system can disable use of carriers
which have interference and poor performance and only use carriers which
meet transmission quality targets.
Further, the ODNs 18 combine downstream video with the telephony
information for transmission onto coaxial distribution legs 30. The video
information from existing video services, generally shown by trunk line
20, is received by and processed by head end 32. Head end 32 or the
central office, includes a video host distribution terminal 34 (VHDT) for
video data interface. The VHDT 34 has optical transmitters associated
therewith for communicating the video information to the remote units 46
via the ODNs 18 of the HFC distribution network
The telephony transmitter 14 of the HDTs 12, shown in FIG. 3 and 4,
includes two transmitters for downstream telephony transmission to protect
the telephony data transmitted. These transmitters are conventional and
relatively inexpensive narrow band laser transmitters. One transmitter is
in standby if the other is functioning properly. Upon detection of a fault
in the operating transmitter, the transmission is switched to the standby
transmitter. In contrast, the transmitter of the VHDT 34 is relatively
expensive as compared to the transmitters of HDT 12 as it is a broad band
analog DFB laser transmitter. Therefore, protection of the video
information, a nonessential service unlike telephony data, is left
unprotected. By splitting the telephony data transmission from the video
data transmission, protection for the telephony data alone can be
achieved. If the video data information and the telephony data were
transmitted over one optical fiber line by an expensive broad band analog
laser, economies may dictate that protection for telephony services may
not be possible. Therefore, separation of such transmission is of
importance.
Further with reference to FIG. 1, the video information is optically
transmitted downstream via optical fiber line 40 to splitter 38 which
splits the optical video signals for transmission on a plurality of
optical fiber lines 42 to a plurality of optical distribution nodes 18.
The telephony transmitter 14 associated with the HDT 12 transmits optical
telephony signals via optical fiber feeder line 42 to the optical
distribution nodes 18. The optical distribution nodes 18 convert the
optical video signals and optical telephony signals for transmission as
electrical outputs via the coaxial distribution portion of the hybrid
fiber coax (HFC) distribution network 11 to a plurality of remote units
46. The electrical downstream video and telephony signals are distributed
to ISUs via a plurality of coaxial legs 30 and coaxial taps 44 of the
coaxial distribution portion of the HFC distribution network 11.
The remote units 46 have associated therewith an ISU 100, shown generally
in FIG. 8, that includes means for transmitting upstream electrical data
signals including telephony information, such as from telephones and data
terminals, and in addition may include means for transmitting set top box
information from set top boxes 45 as described further below. The upstream
electrical data signals are provided by a plurality of ISUs 100 to an
optical distribution node 18 connected thereto via the coaxial portion of
the HFC distribution network 11. The optical distribution node 18 converts
the upstream electrical data signals to an upstream optical data signal
for transmission over an optical fiber feeder line 26 to the head end 32.
FIG. 2 generally shows an alternate embodiment for providing transmission
of optical video and optical telephony signals to the optical distribution
nodes 18 from head end 32, the HDT 12 and VHDT 34 in this embodiment
utilize the same optical transmitter and the same optical fiber feeder
line 36. The signals from HDT 12 and VHDT 34 are combined and transmitted
optically from headend 32 to splitter 38. The combined signal is then
split by splitter 38 and four split signals are provided to the optical
distribution nodes 18 for distribution to the remote units by the coaxial
distribution legs 30 and coaxial taps 44. Return optical telephony signals
from the ODNs 18 would be combined at splitter 38 for provision to the
headend. However, as described above, the optical transmitter utilized
would be relatively expensive due to its broad band capabilities,
lessening the probabilities of being able to afford protection for
essential telephony services.
As one skilled in the art will recognize, the fiber feeder lines 24, 26, as
shown in FIG. 1, may include four fibers, two for transmission downstream
from downstream telephony transmitter 14 and two for transmission upstream
to upstream telephony receiver 16. With the use of directional couplers,
the number of such fibers may be cut in half. In addition, the number of
protection transmitters and fibers utilized may vary as known to one
skilled in the art and any listed number is not limiting to the present
invention as described in the accompanying claims.
The present invention shall now be described in further detail. The first
part of the description shall primarily deal with video sport. The
remainder of the description shall primarily be with regard to telephony
transport.
VIDEO TRANSPORT
The communication system 10 includes the head end 32 which receives video
and telephony information from video and telephony service providers via
trunk line 20. Head end 32 includes a plurality of HDTs 12 and a VHDT 34.
The HDT 12 includes a network interface for communicating telephony
information, such as T1, ISDN, or other data services information, to and
from telephony service providers, such communication also shown generally
by trunk line 20. The VHDT 34 includes a video network interface for
communicating video information, such as cable TV video information and
interactive data of subscribers to and from video service providers, such
communication also shown generally by trunk line 20.
The VHDT 34 transmits downstream optical signals to a splitter 38 via video
optical fiber feeder line 40. The passive optical splitter 38 effectively
makes four copies of the downstream high bandwidth optical video signals.
The duplicated downstream optical video signals are distributed to the
correspondingly connected optical distribution nodes 18. One skilled in
the art will readily recognize that although four copies of the downstream
video signals are created, any number of copies may be made by an
appropriate splitter and that the present invention is not limited to any
specific number.
The splitter is a passive means for splitting broad band optical signals
without the need to employ expensive broad band optical to electrical
conversion hardware. Optical signal splitters are commonly known to one
skilled in the art and available from numerous fiber optic component
manufacturers such as Gould, Inc. In the alternative, active splitters may
also be utilized. In addition, a cascaded chain of passive or active
splitters would further multiply the number of duplicated optical signals
for application to an additional number of optical distribution nodes and
therefore increase further the remote units serviceable by a single head
end. Such alternatives are contemplated in accordance with the present
invention as described by the accompanying claims.
The VHDT 34 can be located in a central office, cable TV head end, or a
remote site and broadcast up to about 112 NTSC channels. The VHDT 34
includes a transmission system like that of a LiteAMp.TM. system available
from American Lightwave Systems, Inc., currently a subsidiary of the
assignee hereof. Video signals are transmitted optically by amplitude
modulation of a 1300 nanometer laser source at the same frequency at which
the signals are received (i.e. the optical transmission is a terahertz
optical carrier which is modulated with the RF video signals). The
downstream video transmission bandwidth is about 54-725 One advantage in
using the same frequency for optical transmission of the video signal as
the frequency of the video signals when received is to provide high
bandwidth transmission with reduced conversion expense. This
same-frequency transmission approach means that the modulation downstream
requires optical to electrical conversion or proportional conversion with
a photodiode and perhaps amplification, but no frequency conversion. In
addition, there is no sample data bandwidth reduction and little loss of
resolution.
An optical distribution node 18, shown in further detail in FIG. 5,
receives the split downstream optical video signal from the splitter 38 on
optical fiber feeder line 42. The downstream optical video signal is
applied to a downstream video receiver 400 of the optical distribution
node 18. The optical video receiver 400 utilized is like that available in
the Lite AMp.TM. product line available from American Lightwave Systems,
Inc. The converted signal from video receiver 400, proportionally
converted utilizing photodiodes, is applied to bridger amplifier 403 along
with converted telephony signals from downstream telephony receiver 402.
The bridger amplifier 403 simultaneously applies four downstream
electrical telephony and video signals to diplex filters 406 which allow
for full duplex operation by separating the transmit and receive functions
when signals of two different frequency bandwidths are utilized for
upstream and downstream transmission. There is no frequency conversion
performed at the ODN 18 with respect to the video or the downstream
telephony signals as the signals are passed through the ODNs to the remote
units via the coaxial portion of the HFC distribution network 11 in the
same frequency bandwidth as they are received at the ODNs 18.
After the ODN 18 has received the downstream optical video signals and such
signals are converted to downstream electrical video signals, the four
outputs of the ODN 18 are applied to four coaxial legs 30 of the coaxial
portion of the HFC distribution network 11 for transmission of the
downstream electrical video signals to the remote units 46. Such
transmission for the electrical video signals occurs in about the 54-725
MHz bandwidth. Each ODN 18 provides for the transmission on a plurality of
coaxial legs 30 and any number of outputs is contemplated in accordance
with the present invention as described in the accompanying claims.
As shown in FIG. 1, each coaxial cable leg 30 can provide a significant
number of remote units 46 with downstream electrical video and telephony
signals through a plurality of coaxial taps 44. Coaxial taps are commonly
known to one skilled in the art and act as passive bidirectional pickoffs
of electrical signals. Each coaxial cable leg 30 may have a number of
coaxial taps 44 connected in series. In addition, the coaxial portion of
the HFC distribution network 11 may use any number of amplifiers to extend
the distance data can be sent over the coaxial portion of such HFC
distribution network 11.
Downstream video signals are provided from the coaxial taps 44 to the
remote units 46. The video signal from the coaxial tap 44 is provided to
an HISU 68 which is generally shown by the block diagram of ISU 100 in
FIG. 8. The ISU 100 is provided with the downstream electrical video and
telephony signal from tap 44 and it is applied to diplex filter 104. The
downstream electrical video and telephony signal is passed through the
diplex filter 104 to both an ingress filter 105 and ISU modem 101. The
downstream video signal is passed by the ingress filter 105 to video
equipment via an optional set top box 45. The downstream electrical
telephony signal applied from the diplex filter 104 to the ISU modem 101
is processed as described in further detail below.
Ingress filter 105 provides the remote unit 46 with protection against
interference of signals applied to the video equipment as opposed to those
provided to other user equipment such as telephones or computer terminals.
Ingress filter 105 passes the video signals; however, it blocks those
frequencies not utilized by the video equipment By blocking those
frequencies not used by the video equipment, stray signals are eliminated
that may interfere with the other services by the network to at least the
same remote unit.
The set top box 45 is an optional element at the remote unit 46.
Interactive video data from set top box 45 would be transmitted by an
additional separate RF modem provided by the video service provider at a
relatively low frequency in the bandwidth of about 5 to 40 MHz. Such
frequency must not be one used for the transport of upstream and
downstream telephony data and downstream video.
For an MISU 66, a separate coaxial line from coaxial tap 44 is utilized to
provide transmission of video signals from the coaxial tap 44 to the set
top box 45 and thus for providing downstream video signals to video
equipment 47. The ingress filter 105 as shown in FIG. 8 is not a part of
the MISU 66 as indicated by its dashed representation.
Alternative embodiments of the VHDT 34 may employ other modulation and
mixing schemes or techniques to shift the video signals in frequency, and
other encoding methods to transmit the information in a coded format. Such
techniques and schemes for transmitting analog video data, in addition to
those transmitting digital video data, are known to one skilled in the art
and are contemplated in accordance with the spirit and scope of the
present invention as described in the accompanying claims.
TELEPHONY TRANSPORT
With reference to FIG. 3, telephony information and ISU operations and
control data (hereinafter referred to as control data) modulated on
carriers by MCC modem 82 is transmitted between the HDT 12 and the
telephony downstream transmitter 14 via coaxial lines 22. Telephony
information and control data modulated on carriers by ISUs 100 is received
at telephony upstream receiver 16 and communicated to the MCC modem 82 via
coaxial cable lines 28. The telephony downstream transmitter 14 and the
telephony upstream receiver 16 transmit and receive, respectively,
telephony information and control data via optical fiber feeder lines 24
and 26 to and from a corresponding optical distribution node 18. The
control data may include all operations, administration, maintenance &
provisioning (OAM&P) for providing the telephony services of the system 10
and any other control data necessary for providing transport of telephony
information between the HDT 12 and the ISUs 100.
A block diagram of the HDT 12 is shown in FIG. 3. The HDT 12 includes the
following modules: Eight DS1 Units (DS1U) (seven quad-DS1 units 48 plus
one protection unit 50), one protection switch & test conversion unit 52
(PSTU), two clock & time slot interchange units 54 (CTSUs) (one active and
one standby/protection unit), six coax master units 56 (CXMUs) (three
active and three standby/protection units), two shelf control units 58
(SCNUs) (one active and one standby/protection unit), and two power supply
units 60 (PWRUs) (two load-sharing units which provide the appropriate HDT
voltages from a central office supply). The DS1U units can also be adapted
to transfer data in the standard E1U format, if desired.
The HDT 12 comprises all the common equipment functions of the telephony
transport of the communication system 10. The HDT 12 is normally located
in a central office and directly interfaces to a local digital switch or
digital network element equipment. The HDT provides the network interface
62 for all telephony information. Each HDT accommodates from 2 to 28 DSX-1
inputs at the network interface 62, representing a maximum of 672 DSO
channels. The HDT 12 also provides all synchronization for telephony
transport in the system 10. The HDT 12 may operate in any one of three
synchronization modes: external timing, line timing or internal timing.
External timing refers to synchronization to a building integrated timing
supply reference which is sourced from a central office in which the HDT
12 is located. Line timing is synchronized to the recovered clock from a
DSX-1 signal normally derived from the local digital switch. Internal
timing is a free-running or hold-over operation where the HDT maintains
its own synchronization in the absence of any valid reference inputs.
The HDT 12 also provides quarter-DS0 grooming capabilities and implements a
4096.times.4096 full-access, non-blocking quarter-DS0 (16 kbps)
cross-connect capability. This allows DS0s and quarter-DS0s (ISDN "D"
channels) to be routed from any timeslot at the DSX-1 network interface 62
to any customer serviced by any ISU 100.
The HDT 12 further provides the RF modem functionality required for
telephony transport over the HFC distribution network 11 including the MCC
modem 82. The HDT 12 accommodates up to three active CXMUs 56 for
providing the modem interface to the HFC distribution network 11 and also
provides one-for-one protection for each active CXMU 56.
The HDT 12 coordinates the telephony transport system including control and
communication of many ISUs of the multi-point to point communication
system 10. Each HDT 12 module performs a function. The DS1U module 48
provides the interface to the digital network and DSX-1 termination. The
PSTU 52 provides DS1U equipment protection by switching the protection
DS1U 50 for a failed DS1U module 48. The CTSU 54 provides the quarter-DS0
timeslot grooming capability and all system synchronization functions. The
CTSU 54 also coordinates all call processing in the system. The CXMU 56,
described in further detail below, provides the modem functionality and
interface for the OFDM telephony transport over the HFC distribution
network 11 and the SCNU 58 supervises the operation of the entire
communication system providing all OAM&P functions for telephony
transport. Most processing of requests for provisioning is performed by
the SCNU 58.
Downstream Telephony Transmitter
The downstream telephony transmitter 14, shown in FIG. 4, takes the coaxial
RF outputs 22 from the active CXMUs 56 of the HDT 12 which carry telephony
information and control data and combines the outputs 22 into a downstream
telephony transmission signal. The electrical-to-optical conversion logic
required for the optical transmission is implemented in a stand-alone
downstream telephony transmitter 14 rather than in the HDT 12 to provide a
more cost effective transport solution By placing this function in a
separate component, the expense of this function does not need to be
replicated in each CXMU 56 of the HDT 12. This reduces the cost of the
CXMU 56 function and allows the CXMU 56 to transmit and receive over coax
instead of fiber. The downstream telephony transmitter 14 also provides
for transmission on redundant downstream fiber feeder lines 24 to an ODN
18.
The downstream telephony transmitter 14 is co-located with the HDT 12
preferably within a distance of 100 feet or less. The downstream telephony
transmitter 14 receives the coaxial RF outputs from the active CXMUs 56,
each within a 6 MHz frequency band, and combines them at combiner 25 into
a single RF signal. Each 6 MHz frequency band is separated by a guard band
as is known to one skilled in the art. Downstream telephony information is
then transmitted in about the 725-800 MHz frequency band. The telephony
transmitter 14 passes the combined signal through a 1-to-2 splitter (not
shown), thereby producing redundant downstream electrical signals. The two
redundant signals are each delivered to redundant laser transmitters 501
for electrical-to-optical conversion and the redundant signals modulate an
optical output such that the output of the downstream telephony
transmitter 14 is on two optical feeder lines 24, each having an identical
signal modulated thereon. This provides protection for the downstream
telephony portion of the present system. Both Fabry-Perot lasers in the
telephony transmitter 14 are active at all times. AU protection functions
are provided at the receive end of the optical transmission (located at
the ODN 18) where one of two receivers is selected as "active;" therefore,
the telephony transmitter 14 requires no protection switching
capabilities.
Upstream Telephony Receiver
The upstream telephony receiver 16 performs the optical-to-electrical
conversion on the upstream optical telephony signals on the upstream
optical feeder lines 26 from the ODN 18. The upstream telephony receiver
16 is normally co-located in the central office with the HDT 12, and
provides an electrical coaxial output to the HDT 12, and a coaxial output
23 to be provided to a video set-top controller (not shown). Upstream
telephony information is routed via coax lines 28 from the upstream
telephony receiver 16 to active CXMUs 56 of the HDT 12. The coaxial link
28 between the HDT 12 and the upstream telephony receiver 16 is preferably
limited to a distance of 100 feet or less and is an intra-office link.
Video set-top controller information, as described in the Video Transport
section hereof, is located in a bandwidth of the RF spectrum of 5-40 MHz
which is not utilized for upstream telephony transport such that it is
transmitted along with the upstream telephony information.
The upstream telephony receiver 16 has dual receivers 502 for the dual
upstream optical fiber feeders lines 26. These feeder lines 26 carry
redundant signals from the ODN 18 which contain both telephony information
and control data and also video set-top box information. The upstream
telephony receiver 16 performs automatic protection switching on the
upstream feeder lines 26 from the ODN. The receiver 502 selected as
"active" by protection logic is split to feed the coaxial outputs 28 which
drive the HDT 12 and output 23 is provided to the set-top controller (not
shown).
Optical Distribution Node
Referring to FIG. 5, the ODN 18 provides the interface between the optical
feeder lines 24 and 26 from the HDT 12 and the coaxial portion of the HFC
distribution network 11 to the remote units 46. As such, the ODN 18 is
essentially an optical-to-electrical and electrical-to-optical converter.
The maximum distance over coax of any ISU 100 from an ODN 18 is preferably
about 6 km and the maximum length of the combined optical feeder
line/coaxial drop is preferably about 20 km. The optical feeder line side
of the ODN 18 terminates six fibers although such number may vary. They
include: a downstream video feeder line 42 (single fiber from video
splitter 38), a downstream telephony feeder line 24 (from downstream
telephony transmitter 14), a downstream telephony protection feeder line
24 (from downstream telephony transmitter 14), an upstream telephony
feeder line 26 (to upstream telephony receiver 16), an upstream protection
feeder line 26 (to upstream telephony receiver 16), and a spare fiber (not
shown). The ODN 18 provides protection switching functionality on the
receive optical feeder lines 24 from the downstream telephony transmitter.
The ODN provides redundant transmission on the upstream optical feeder
lines 26 to the upstream telephony receiver. Protection on the upstream
optical feeder lines is controlled at the upstream telephony receiver 16.
On the coaxial distribution side of ODN 18, the ODN 18 terminates up to
four coaxial legs 30.
In the downstream direction, the ODN 18 includes downstream telephony
receiver 402 for converting the optical downstream telephony signal into
an electrical signal and a bridger amplifier 403 that combines it with the
converted downstream video signal from downstream video receiver 400
terminated at the ODN 18 from the VHDT 34. This combined wide-band
electrical telephony/video signal is then transported in the spectrum
allocated for downstream transmission, for example, the 725-800 MHz band,
on each of the four coaxial legs of the coaxial portion of the HFC
distribution network 11. As such, this electrical telephony and video
signal is carried over the coaxial legs 30 to the ISUs 100; the bridger
amplifier 403 simultaneously applying four downstream electrical telephony
and video signals to diplex filters 406. The diplex filters 406 allow for
full duplex operation by separating the transmit and receive functions
when signals at two different frequency bandwidths are utilized for
upstream and downstream transmission. There is no frequency conversion
available at the ODN 18 for downstream transport as the telephony and
video signals are passed through the ODN 18 to the remote units 46 via the
coaxial portion of HFC distribution network 11 in the same frequency
bandwidth as they are received at the ODN 18. As shown in FIG. 1, each
coaxial leg 30 can provide a significant number of remote units 46 with
downstream electrical video and telephony signals through a plurality of
coaxial taps 44. Coaxial taps 44 commonly known to one skilled in the art
act as passive bidirectional pickoffs of electrical signals. Each coaxial
leg 30 may have a number of coaxial taps connected in a series. In
addition, the coaxial portion of the HFC distribution network 11 may use
any number of amplifiers to extend the distance data can be sent over the
coaxial portions of the system 10. The downstream electrical video and
telephony signals are then provided to an ISU 100 (FIG. 8), which, more
specifically, may be an HISU 68 or an MISU 66 as shown in FIG. 1.
In the upstream direction, telephony and set top box information is
received by the ODN 18 at diplex filters 406 over the four coaxial legs 30
in the RF spectrum region from 5 to 40 MHz. The ODN 18 may include
optional frequency shifters 64 equipped on up to three of four coaxial
legs 30. These frequency shifters 64, if utilized, mix the upstream
spectrum on a coaxial leg to a higher frequency prior to combining with
the other three coaxial legs. Frequency shifters 64 are designed to shift
the upstream spectrum in multiples of 50 MHz. For example, the frequency
shifters 64 may be provisioned to mix the upstream information in the 5-40
MHz portion of the RF spectrum to any of the following ranges: 50 to 100
MHz, 100 to 150 MHz, or 150 to 200 MHz. This allows any coaxial leg 30 to
use the same portion of the upstream RF spectrum as another leg without
any spectrum contention when the upstream information is combined at the
ODN 18. Provisioning of frequency shifters is optional on a coaxial leg
30. The ODN 18 includes combiner 408 which combines the electrical
upstream telephony and set top box information from all the coaxial legs
30 (which may or may not be frequency shifted) to form one composite
upstream signal having all upstream information present on each of the
four coaxial legs 30. The composite electrical upstream signal is
passively 1:2 split and each signal feeds an upstream Fabry-Perot laser
transmitter which drives a corresponding upstream fiber feeder line 26 for
transmission to the upstream telephony receiver 16.
FIG. 6 illustrates an embodiment of a frequency shifter, indicated
generally at 64', for use in ODN 18 of FIG. 5. Frequency shifter 64'
comprises a mixer 700 that is coupled to receive and shift the frequency
band of RF signals in the upstream direction from diplex filter 406 for a
coaxial leg 30. An output of mixer 700 is coupled through a bandpass
filter 704 to combiner 408. Local oscillator 702 is coupled to provide a
signal to control the operation of mixer 700.
In operation, frequency shifter 64' shifts a block of RF signals from a
first frequency range to a second frequency range. For example, as
mentioned above, the RF signals provided to frequency shifter may comprise
RF signals in the range from 5 to 40 Mhz. In one embodiment, ODN 18
comprises three frequency shifters 64'. In this embodiment, the local
oscillators 702 of the three frequency shifters provide signals of 76 MHZ,
149 MHZ, and 222 MHZ, respectively. Thus, frequency shifters 64'
respectively shift the upstream RF signals approximately to the 50 to 100
MHZ, 125 to 175 MHZ and 200 to 250 MHZ ranges.
If the upstream telephony and set top box signals are upshifted at the ODN
18, the upstream telephony receiver 16 includes frequency shifters 31 to
downshift the signals according to the upshifting done at the ODN 18. A
combiner 33 then combines the downshifted signals for application of a
combined signal to the HDT 12. Such downshifting and combining is only
utilized if the signals are upshifted at the ODN 18.
FIG. 7 illustrates an embodiment of a frequency shifter, indicated
generally at 31' for use in telephony upstream receiver 16 of FIG. 8.
Frequency shifter 31' returns a block of RF signals shifted by frequency
shifter 64' to original frequency range of the block. For example,
frequency shifter 31' may return a block of RF signals to 5 to 40 MHZ from
50 to 100 MHZ.
As discussed in more detail below, the upstream telephony signals processed
by frequency shifters 31' and 64' are typically OFDM signals. Thus,
frequency shifters 64' must return the RF signals to the original
frequency range without introducing adverse phase and frequency errors. To
reduce the likelihood of this corruption of the OFDM signals, frequency
shifter 31' locks its local oscillator to the local oscillator of a
corresponding frequency shifter 64' using a pilot tone transmitted from
ODN 18 to telephony upstream receiver 16.
Frequency shifter 31' includes a bandpass filter 706 that is coupled to
receive an RF signal from ODN 18. Bandpass filter 706 is coupled to a
splitter 708. Splitter 708 is coupled to provide the RF signal to an input
of mixer 718. Further, splitter 708 provides a second output that is used
to generate a local oscillator signal for mixer 718. This local oscillator
signal is phase locked with a corresponding local oscillator 702 of
frequency converter 64'. This second output of splitter 708 is coupled to
phase detector 712 through bandpass filter 710. Phase detector 712 is
coupled to provide a control signal to voltage controlled oscillator 714.
Voltage controlled oscillator 714 is coupled through splitter 716 to
provide the local oscillator signal to mixer 718. Splitter 716 further
provides a feedback signal to phase detector 712.
In operation, phase detector 712 phase locks local oscillator signal of
frequency shifter 31' with local oscillator 702 of a corresponding
frequency shifter 64'. Phase detector 712 compares the pilot tone from ODN
18 with the feedback signal from voltage controlled oscillator 714 to
generate the control signal for voltage controlled oscillator 714.
Consequently, the local oscillator signal provided to mixer 718 is phase
locked with the corresponding local oscillator 702 of frequency shifter
64'. Mixer 718 uses the local oscillator signal from splitter 716 and
voltage controlled oscillator 714 to shift the block of RF signals
received by frequency shifter 31' to the original frequency range of the
block of RF signals. Advantageously, unacceptable modifications of the
OFDM upstream signal by frequency shifters 64' and 31' are thus avoided.
Integrated Services Unit (ISUs)
Referring to FIG. 1, the ISUs 100, such as HISU 68 and MISU 66, provide the
interface between the HFC distribution network 11 and the customer
services for remote units 46. Two basic types of ISUs are shown, which
provide service to specific customers. Multiple user integrated service
unit 66 (MISUs) may be a multiple dwelling integrated service unit or a
business integrated service unit. The multiple dwelling integrated service
unit may be used for mixed residential and business environments, such as
multi-tenant buildings, small businesses and clusters of homes. These
customers require services such as plain old telephone service (POTS),
data services, DS1 services, and standard TR-57 services. Business
integrated service units are designed to service business environments.
They may require more services, for example, data services, ISDN, DS1
services, higher band-width services, such as video conferencing, etc.
Home integrated services units 68 (HISUs) are used for residential
environments such as single-tenant buildings and duplexes, where the
intended services are POTS and basic rate integrated digital services
network (ISDN). Description for ISUs shall be limited to the HISUs and
MISUs for simplicity purposes as multiple dwelling and business integrated
service units have similar functionality as far as the present invention
is concerned.
All ISUs 100 implement RF modem functionality and can be generically shown
by ISU 100 of FIG. 8. ISU 100 includes ISU modem 101, coax slave
controller unit (CXSU) 102, channel units 103 for providing customer
service interface, and diplex filter/tap 104. In the downstream direction,
the electrical downstream telephony and video signal is applied to diplex
filter/tap 104 which passes telephony information to ISU modem 101 and
video information to video equipment via an ingress filter 105 in the case
of a HISU. When the ISU 100 is a MISU 66, the video information is
rejected by the diplex filter. The ISU modem 101 demodulates the
downstream telephony information utilizing a modem corresponding to the
MCC modem 82 used for modulating such information on orthogonal
multicarriers at HDT 12. ISU 100 demodulates downstream telephony
information from a coaxial distribution leg 30 in a provisionable 6 MHz
frequency band. Timing generation 107 of the ISU modem 101 provides
clocking for CXSU 102 which provides processing and controls reception and
transmission by ISU modem 101. The demodulated data from ISU modem 101 is
passed to the applicable channel units 103 via CXSU 102 depending upon the
service provided. For example, the channel units 103 may include line
cards for POTS, DS1 services, ISDN, other data services, etc. Each ISU 100
provides access to a fixed subset of all channels available in a 6 MHz
frequency band corresponding to one of the CXMUs of HDT 12. This subset of
channels varies depending upon the type of ISU 100. An MISU 66 may provide
access to many DS0 channels in a 6 MHz frequency band, while an HISU 68
may only provide access to a few DS0 channels.
The channel units 103 provide telephony information and control data to the
CXSU 102, which provides such data to ISU modem 101 and controls ISU modem
101 for modulation of such telephony data and control data in a
provisional 6 MHz frequency band for transmission onto the coaxial
distribution leg 30 connected thereto. The upstream 6 MHz frequency band
provisionable for transmission by the ISU 100 to the HDT 12 corresponds to
one of the downstream 6 MHz bands utilized for transmission by the CXMUs
56 of HDT 12.
The CXSU 102 which applies demodulated data from the ISU modem 101 to the
applicable channel units, performs data integrity checking on the
downstream 10 bit DS0+ packets received from the ISU modem 101. Each ten
bit DS0+ packet as described below includes a parity or data integrity
bit. The CXSU 102 will check the parity of each downstream 10 bit DS0+
channel it receives. Further, the parity of each upstream DS0+ received
from the channel units 103 is calculated and a parity bit inserted as the
tenth bit of the upstream DS0+ for decoding and identification by the HDT
12 of an error in the upstream data. If an error is detected by CXSU 102
when checking the parity of a downstream 10 bit DS0+ channel it receives,
the parity bit of the corresponding upstream channel will be intentionally
inverted to inform the HDT 12 of a parity error in the downstream
direction. Therefore, the upstream parity bit is indicative of errors in
the downstream DS0+ channel and the corresponding upstream DS0+ channel.
An example of such a parity bit generation process is described in U.S.
patent application 08/074,913 entitled "Point-to Multipoint Performance
Monitoring and Failure Isolation System" assigned to the assignee hereof
and entirely incorporated herein by reference. This upstream parity bit is
utilized in channel monitoring as described further below. As would be
apparent to one skilled in the art, the parity checking and generation may
be performed, at least in part, in other elements of the ISU or associated
therewith such as the channel units.
Each ISU 100 recovers synchronization from downstream transmission,
generates all clocks required for ISU data transport and locks these
clocks to the associated HDT timing. The ISUs 100 also provide call
processing functionality necessary to detect customer line seizure and
line idle conditions and transmit these indications to the HDT 12. ISUs
100 terminate and receive control data from the HDT 12 and process the
control data received therefrom. Included in this processing are messages
to coordinate dynamic channel allocation in the communication system 10.
Finally, ISUs 100 generate ISU operating voltages from a power signal
received over the HFC distribution network 11 as shown by the power signal
100 taken from diplex filter/tap 104.
Data Path in HDT
The following is a detailed discussion of the data path in the host digital
terminal (HDT) 12. Referring to FIG. 3, the data path between the network
facility at the network interface 62 and the downstream telephony
transmitter 14 proceeds through the DS1U 48, CTSU 54, and CXMU 56 modules
of the HDT 12, respectively, in the downstream direction. Each DS1U 48 in
the HDT 12 takes four DS1s from the network and formats this information
into four 24-channel, 2.56 Mbps data steams of modified DS0 signals
referred to as CTSU inputs 76. Each DS0 in the CTSU input has been
modified by appending a ninth bit which can carry multiframe timing,
signaling information and control/status messages (FIG. 9). This modified
DS0 is referred to as a "DS0+." The ninth bit signal (NBS) carries a
pattern which is updated each frame and repeats every 24 frames. This maps
each 64 kbps DS0 from the network into a 72 kbps DS0+. Thus, the
twenty-four DS0 channels available on each DS1 are formatted along with
overhead information into twenty-four DS0+ channels on each of four CTSU
input streams.
The ninth bit signaling (NBS) is a mechanism developed to carry the
multiframe timing, out-of-band signaling bits and miscellaneous status and
control information associated with each DS0 between the DS1U and the
channel units. Its main functions are to carry the signaling bits to
channel units 103 and to provide a multiframe clock to the channel units
103 so that they can insert upstream bit signaling into the DS0 in the
correct frame of the multiframe. Because downstream DS0 s may be coming
from DS1s which do not share the same multiframe phase each DS0 must carry
a multiframe clock or marker which indicates the signaling frames
associated with the origination DS1. The NBS provides this capability.
Ninth bit signaling is transparent to the OFDM modem transport of the
communication system 10.
Up to eight DS1Us 48 may be equipped in a single HDT 12; including seven
active DS1Us 48 and a protection DS1U module 50. Thus, 32 CTSU inputs are
connected between the DS1Us and the CTSUs 54 but a maximum of 28 can be
enabled to carry traffic at any one time. The four remaining CTSU inputs
are from either the protection DS1U or a failed DS1U. The PSTU includes
switch control for switching the protection DS1U 50 for a failed DS1U.
Each CTSU input is capable of carrying up to 32, 10-bit channels, the first
24 channels carry DS0+s and the remaining bandwidth is unused. Each CTSU
input 76 is clocked at 2.56 Mbps and is synchronized to the 8 kHz internal
frame signal (FIG. 11). This corresponds to 320 bits per 125 .mu.sec frame
period. These 320 bits are framed as shown in FIG. 9. The fourteen gap
bits 72 at the beginning of the frame carry only a single activity pulse
in the 2nd bit position, the remaining 13 bits are not used. Of the
following 288 bits, the first 216 bits normally carry twenty-four DS0+
channels where each DS0+ corresponds to a standard 64 kbps DS0 channel
plus the additional 8 kbps signaling bit. Thus, each DS0 + has a bandwidth
of 72 kbps (nine bits every 8 kHz frame). The remaining 72 bits are
reserved for additional DS0+ payload channels. The final eighteen bits 74
of the frame are unused gap bits.
The clock and time slot interchange unit 54 (CTSU) of the HDT 12 takes
information from up to 28 active CTSU input data streams 76 and
cross-connects them to up to twenty-four 32-channel, 2.56 Mbps output data
streams 78 which are input to the coax master units (CXMUs) 56 of the HDT
12. The format of the data streams between the CTSU 54 and the CXMUs 56 is
referred to as a CTSU output. Each CTSU output can also carry up to 32,
10-bit channels like the CTSU input. The first 28 carry traffic and the
remaining bandwidth is unused. Each CTSU output is clocked at 2.56 Mbps
and is synchronized to the 8 kHz internal framing signal of the HDT 12
(FIG. 11). This corresponds to 320 bits per 125 .mu.sec frame period. The
frame structure for the 320 bits are as described above for the CTSU input
structure.
The HDT 12 has the capability of time and space manipulation of quarter-DS0
packets (16 kbps). This function is implemented with the time slot
interchange logic that is part of CTSU 54. The CTSU implements a
4096.times.4096 quarter-DS0 cross-connect function, although not all time
slots are utilized In normal operation, the CTSU 54 combines and relocates
up to 672 downstream DS0+ packets (or up to 2688 quarter-DS0 packets)
arranged as 28 CTSU inputs of 24 DS0+s each, into 720 DS0+ packets (or
2880 quarter-DS0 packets) arranged as 24 CTSU outputs of 32 DS0+s each.
The system has a maximum throughput of 672 DS0+ packets at the network
interface so not all of the CTSU output bandwidth is usable. If more than
the 672 channels are assigned on the "CTSU output" side of the CTSU, this
implies concentration is being utilized. Concentration is discussed
further below.
Each CXMU 56 is connected to receive eight active CTSU outputs 78 from the
active CTSU 54. The eight CTSU outputs are clocked by a 2.56 MHz system
clock and each carries up to 32 DS0+s as described above. The DS0+s are
further processed by the CXMU 56 and a tenth parity bit is appended to
each DS0+ resulting in a 10 bit DS0+. These 10 bit packets contain the
DS0, the NBS (ninth bit signal) and the parity or data integrity bit (FIG.
10). The 10 bit packets are the data transmitted on the HFC distribution
network 11 to the ISUs 100. The 10th bit or data integrity bit inserted in
the downstream channels is decoded and checked at the ISU and utilized to
calculate and generate a parity bit for corresponding channels in the
upstream as described above. This upstream parity bit which may be
representative of an error in the downstream or upstream channel is
utilized to provide channel protection or monitoring as further described
herein.
CTSU 54 is also capable of applying a conventional Reed-Soloman code to
transmitted payload data for running error correction. Such codes carry an
overhead which must be borne by the inclusion of error-correcting symbols
in each block or message transmitted. Generally, two check symbols are
required to correct one corrupted data symbol in a message. (The incorrect
symbol may contain any number of errors in its 5 bits, as long as all bit
errors are confined to the same symbol. But even a single incorrect bit in
each of two symbols counts as two errors.) Short messages impose less
computational overhead on a system, but can correct fewer errors in the
message. Conversely, long messages require more computation and more
latency before the corrections can be applied, but their error-correction
ability is greater. FIG. 54 represents, for an example system, the
probability of an uncorrectable error in a frame for various error
probabilities in one individual symbol. The solid curve shows the error
performance for a 21-frame message having 19 frames of data symbols and
two frames of error-correction code; the dashed curve represents a
41-frame message having 37 data and four code frames; the dotted curve
gives the best performance, with 73 data frames and eight code frames in
an 81-frame message.
The present system allows a choice of different error-correction abilities
for different types of data. For example, voice data is highly redundant,
and needs little defense against errors. Financial transaction data, on
the other hand, wants a large degree of data integrity. In addition, it
may be desirable to allow a user to select--and pay for--whatever degree
of error correction that he desires. CTSU 54, FIG. 3, includes a
conventional "provisioning table", which specifies a number of parameters
relating to particular payload channels. FIG. 55 shows a provisioning
table 4411 having an added column containing indications for several
different amounts of error protection. In method 4410, step 4412 reads the
entry for a particular channel to be set up. In this implementation, the
entry may specify message lengths of 21, 41, or 81 bits, respectively
having the ability to correct 1, 2, or 4 symbols; the entry may also
specify no correction, in which case message blocks do not apply. Step
4413 encodes the table entry in an IOC message and sends it to the ISU
whose address appears in that row of table 4111. A general-purpose
processor in CXSU 102 of the ISU stores the frame length in step 4414. As
the CXSU receives data from modem 101, FIG. 8, it decodes the frames of an
entire message, 4415, then decodes the check symbols for the message,
4416, and signals an error, 4417, if one exists in the message. Steps
4415-4417 repeat for subsequent messages. The ISU employs the same process
to send frames upstream to the head end, using the frame length setting
specified in step 4414.
Within both CXSU 102 at the ISU and the CXMU, FIG. 12 at the HDT, a
21-frame message or block requires 19 symbol or frame times to decode the
message, then has two frames of latency while its two check symbols are
decoded. A 41 -frame message uses four frames of time for computation of
any errors from the four check symbols following its 37 data symbols. An
81-frame message presents any error indication 8 frame times after the end
of its 73 data frames. (One extra frame of delay is imposed in the
downstream direction due to remapping at the HDT.) If all messages were to
start at the same time for all channels in an entire band, the
computational load in the HDT would peak during the check-symbol frames,
and would be lower at other times. Since the processor must be capable of
handling the peak loads, its power is underutilized at other times.
The present system allows a lower-capacity processor to handle error
correction by staggering the beginning times of different messages in
different subbands of channels, so that not all of them come due at the
same time. That is, the start of a message in any channel of a subband is
offset from the start of a message or "multiframe" signal, to be
described, by a predetermined number of frame times of 125 .mu.sec. each.
The table below shows how the 24 subbands of FIG. 16 are offset, for each
message length which can be selected.
TABLE 1
Subband 21 frames 41 frames 81 frames
0 0 0 0
1 0 0 1
2 0 1 2
3 0 1 3
4 1 2 4
5 1 2 5
6 1 3 6
7 1 3 7
8 2 4 8
9 2 4 9
10 2 5 10
11 2 5 11
12 3 6 12
Only 13 subband settings are required, since no unit tunes more than 130
channels. Giving all 10 channels of each subband the same offset does not
overload the processors of the remote units. However, the head end (HDT),
which receives and transmits all channels, can enjoy significant relief
from not having to encode or decode the check symbols for all channels at
the same time.
FIG. 56 shows steps 4120 for performing frame staggering. Step 4421 repeats
method 4420 for all active payload channels. Step 4422 accesses the
current messages for the channels in one subband? Step 4423 calculates the
1, 2, or 4 Reed-Soloman check words for the 21, 41, or 81 message data
words. Step 4424 waits N frames past the start of a multiframe, whereupon
step 4425 sends the message to modem 82, FIG. 3 for transmission.
At a remote ISU, CXSU performs the same steps 4420 for upstream messages.
Step 4422 is required only in an MISU, because all channels in an HISU
reside in the same subband. Method 4420 may be performed at both ends of
the system, as described herein; it may also be performed only at one end,
either HDT or ISU. Staggering from the ISU to the HDT is preferable if
only one end is staggered, because the most critical processing load is
the error-correction of all 240 channels in the upstream receiving modem,
FIG. 26.
The use of error-correcting codes along with unencoded data raises problems
in a real-time transport system. Data arrives from the trunk line 20, FIG.
1, at a constant rate. This data must be transmitted downstream in the
same time duration, whether it is encoded along the way, or sent
unencoded. Likewise, upstream data usually must be transmitted at the same
rate whether or not it is encoded. That is, the use of error-correcting
codes must be time-transparent at both ends of the system. But
error-correcting codes require the transmission of check digits or symbols
along with the data. The present system resolves this difficulty by
packing the data words differently if they are encoded. As explained
above, the basic unencoded word length for a DS0++ channel is ten bits:
eight data bits, a signaling (NBS) bit, and a parity bit. When encoding is
used, however, this format is changed to nine-bit words, with a single
parity bit for the entire message. This is the reason for the choice of
frame sizes for the encoded modes. A 21 -frame message contains 19 data
frames, which would ordinarily be transmitted as 10.times.19=190 bits.
Those same data frames, packaged as nine-bit words along with two nine-bit
check words, require (19+2).times.9=189 bits; adding one more parity bit
covering the entire message lock gives 190 bits, the same number as that
required for the unencoded version of the same data. The 41-frame message
has 37 frames of data, or 370 bits in unencoded 10-bit format. Encoded as
37 nine-bit words along with four check words, the same message requires
(37+4).times.9=369 bits; again, a single additional parity bit yield the
370 bits of the same data in unencoded form. The 81-frame format has 73
data words, 8 check words, and a parity bit, yielding the same number of
bits as 73 data words in 10-bit form.
There are many other combinations of numbers which yield similar results.
These can be found heuristically without a great deal of experimentation.
The first step is to estimate rough numbers of large (parity-bearing)
words in one or more message sizes, and the number of errors desired to be
correctable for each size. The next step is to determine a number of
smaller (non-parity) words that carry the same amount of data, but which
form a total message the same size or slightly smaller than the total
number of bits in the large-word format. Any excess bits then are assigned
to parity over the block--or to any other function, for that matter. For
example, if two bits are left over instead of one for each message, they
could represent two parity bits over the message, two control or
format-designation bits, etc. The use of check symbols, of course, greatly
reduces the need for parity or other forms of error detection. In fact,
while the present system uses the message-parity bit as parity in the
downstream direction, the ISU deliberately sets the parity bit to an
incorrect value in an upstream message if it was incorrect in the
downstream message. This serves to signal the HDT that a bit error was
encountered, when the HDT would not otherwise be aware of it; this in turn
allows the HDT to keep more accurate statistics on channel quality for
reallocating channels, or for other purposes.
FIG. 57 shows a method 4430 for adding the "code packing" feature to the
method 4420 of FIG. 56. Step 4431 repeats the steps for all channels. Step
4432 determines whither the data for the channel is to be encoded or not.
If not, step 4433 merely transmits it word-by-word to the modem. If it is
to be encoded, step 4434 strips the parity (or other) bit(s) from each
word. After step 4435 has formed the check words, step 4436 calculates the
message-wide parity, or other desired function. Thereafter, step 4437
waits the proper number of frame times (as specified by method 4420, and
step 4338 sends the message to the modem as before.
In the upstream direction, the reverse path through the HDT is
substantially a mirror of the forward path through the HDT 12. For
example, the tenth parity bit is processed at the CXMU 56 and the signal
from the CXMU 56 to the CTSU 54 is in the format of FIG. 9.
The round trip delay of a DS0 is the same for every data path. The time
delay over the path from the downstream CTSU output, through CXMU 56, over
the HFC distribution network to the ISU 100 and then from the ISU 100,
back over the HFC distribution network 11 through CXMU 56 and to CTSU 54
is controlled by upstream synchronization, as described in detail below.
Generally, path delay is measured for each ISU and if it is not the
correct number of frames long, the delay length is adjusted by adding
delay to the path at the ISU 100.
Coax Master Unit (CXMU)
The coax master unit 56 (CXMU), shown in FIG. 3, includes the coax master
card logic 80 (CXMC) and the master coax card (MCC) modem 82. As
previously described, up to six CXMUs may be equipped in an HDT 12. The 6
CXMUs 56 include three pairs of CXMUs 56 with each pair providing for
transmit in a 6 MHz bandwidth. Each pair of CXMUs 56 includes one active
CXMU and a standby CXMU. Thus, one to one protection for each CXMU is
provided. As shown in FIG. 3, both CXMUs of the pair are provided with
upstream telephony data from the upstream telephony receiver 16 and are
capable of transmitting via the coaxial line 22 to the downstream
telephony transmitter 14. As such, only a control signal is required to
provide for the one-to-one protection indicating which CXMU 56 of the pair
is to be used for transmission or reception.
Coax Master Card Logic (CXMC)
The coax master card logic 80 (CXMC) of the CXMU 56 (FIG. 12), provides the
interface between the data signals of the HDT 12, in particular of the
CTSU 54, and the modem interface for transport of data over the HFC
distribution network 11. The CXMC 80 interfaces directly to the MCC modem
82. The CXMC 80 also implements an ISU operations channel transceiver for
multi-point to point operation between the HDT 12 and all ISUs 100
serviced in the 6 MHz bandwidth in which the CXMU 56 controls transport of
data within. Referring to FIG. 12, the CXMC includes controller and logic
84, downstream data conversion 88, upstream data conversion 90, data
integrity 92, IOC transceiver 96, and timing generator 94.
Downstream data conversion 88 performs the conversion from the nine-bit
channel format from CTSU 54 (FIG. 9) to the ten-bit channel format (FIG.
10) and generates the data integrity bit in each downstream channel
transported over the HFC distribution network 11. The data integrity bit
represents odd parity. Downstream data conversion 88 is comprised of at
least a FIFO buffer used to remove the 32 gap bits 72, 74 (FIG. 9) present
in the downstream CTSU outputs and insert the tenth, data integrity bit,
on each channel under control of controller and logic 84.
The upstream data conversion 90 includes at least a FIFO buffer which
evaluates the tenth bit (data integrity) appended to each of the upstream
channels and passes this information to the data integrity circuitry 92.
The upstream data conversion 90 converts the data stream often-bit
channels (FIG. 10) back to the nine-bit channel format (FIG. 9) for
application to CTSU 54. Such conversion is performed under control of
controller and logic 84.
The controller and logic 84 also manages call processing and channel
allocation for the telephony transport over the HFC distribution network
11 and maintains traffic statistics over the HFC distribution network 11
in modes where dynamic time-slot allocation is utilized, such as for
providing TR-303 services, concentration services commonly known to those
skilled in the art In addition, the controller 84 maintains error
statistics for the channels in the 6 MHz band in which the CXMU transports
data, provides software protocol for all ISU operations channel
communications, and provides control for the corresponding MCC modem 82.
The data integrity 92 circuitry processes the output of the tenth bit
evaluation of each upstream channel by the upstream conversion circuit 90.
In the present system, parity is only guaranteed to be valid on a
provisioned channel which has a call in progress. Because initialized and
activated ISU transmitters may be powered down when the ISUs are idle, the
parity evaluation performed by the CXMC is not always valid. A parity
error detected indicates either a transmission error in an upstream
channel or a transmission error in a downstream channel corresponding to
the upstream channel.
The ISU operations channel (IOC) transceiver 96 of the CXMC 80 contains
transmit buffers to hold messages or control data from the controller and
logic 84 and loads these IOC control messages which are a fixed total of 8
bytes in length into a 64 kbps channel to be provided to the MCC modem 82
for transport on the HFC distribution network 11. In this implementation,
all IOC channels carry the same information at all times. That is, the IOC
messages are broadcast simultaneously over all the channels. This allows
the use of inexpensive and rugged narrow-band modems in the ISUs,
reserving the more expensive and critical wideband models for the HDT,
which uses only one modem for an entire 6 MHz band, and which can be
located centrally in a controlled environment. In the upstream direction,
the IOC transceiver receives the 64 kbps channel via the MCC modem 82
which provides the controller and logic 84 with such messages.
The timing generator circuit 94 receives redundant system clock inputs from
both the active and protection CTSUs 54 of the HDT 12. Such clocks include
a 2 kHz HFC multiframe signal, which is generated by the CTSU 54 to
synchronize the round trip delay on all the coaxial legs of the HFC
distribution network. This signal indicates multiframe alignment on the
ISU operations channel and is used to synchronize symbol timing and data
reconstruction for the transport system. A 8 kHz frame signal is provided
for indicating the first "gap" bit of a 2.56 MHz, 32 channel signal from
the CTSU 54 to the CXMU 56. A 2.048 MHz clock is generated by the CTSU 54
to the SCNU 58 and the CXMU 56. The CXMU 56 uses this clock for ISU
operations channel and modem communication between the CXMC 80 and the MCC
modem 82. A 2.56 MHz bit clock is used for transfer of data signals
between the DS1Us 48 and CTSUs 54 and the CTSUs 54 and CXMCs 56. A 20.48
MHz bit clock is utilized for transfer of the 10-bit data channels between
the CXMC and the MCC.
Master Coax Card (MCC) Modem
The master coax card (MCC) modem 82 of the CXMU 56 interfaces on one side
to the CXMC 80 and on the other side to the telephony transmitter 14 and
receiver 16 for transmission on and reception from the HFC distribution
network 11. The MCC modem 82 implements the modem functionality for OFDM
transport of telephony data and control data. The block diagram of FIG. 3
identifies the associated interconnects of the MCC modem 82 for both
upstream and downstream communication. The MCC modem 82 is not an
independent module in the HDT 12, as it has no interface to the HDT 12
other than through the CXMC 80 of the CXMU 56. The MCC modem 82 represents
the transport system logic of the HDT 12. As such, it is responsible for
implementing all requirements associated with information transport over
the HFC distribution network 11. Each MCC modem 82 of the CXMUs 56 of HDT
12 is allocated a maximum bandwidth of 6 MHz in the downstream spectrum
for telephony data and control data transport. The exact location of the 6
MHz band is provisionable by the CXMC 80 over the communication interface
via the IOC transceiver 96 between the CXMC 80 and MCC modem 82. The
downstream transmission of telephony and control data is in the RF
spectrum of about 725 to 800 MHz.
Each MCC modem 82 is allocated a maximum of 6 MHz in the upstream spectrum
for receipt of control data and telephony data from the ISUs within the RF
spectrum of about 5 to 40 MHz Again, the exact location of the 6 MHz band
is provisionable by the CXMC 80 over the communication interface between
the CXMC 80 and the MCC modem 82.
The MCC modem 82 receives 256 DS0+ channels from the CXMC 80 in the form of
a 20.48 MHz signal as described previously above. The MCC modem 82
transmits this information to all the ISUs 100 using the multicarrier
modulation technique based on OFDM as previously discussed herein. The MCC
modem 82 also recovers 256 DS0+ multicarrier channels in the upstream
transmission over the HFC distribution network and converts this
information into a 20.48 Mbps stream which is passed to CXMC 80. As
described previously, the multicarrier modulation technique involves
encoding the telephony and control data, such as by quadrature amplitude
modulation, into symbols, and then performing an inverse fast Fourier
transform technique to modulate the telephony and control data on a set of
orthogonal multicarriers.
Symbol alignment is a necessary requirement for the multicarrier modulation
technique implemented by the MCC modem 82 and the ISU modems 101 in the
ISUs 100. In the downstream direction of transmission, all information at
an ISU 100 is generated by a single CXMU 56, so the symbols modulated on
each multicarrier are automatically phase aligned. However, upstream
symbol alignment at a receiver of the MCC modem 82 varies due to the
multi-point to point nature of the HFC distribution network 11 and the
unequal delay paths of the ISUs 100. In order to maximize receiver
efficiency at the MCC modem 82, all upstream symbols must be aligned
within a narrow phase margin. This is done by utilizing an adjustable
delay parameter in each ISU 100 such that the symbol periods of all
channels received upstream from the different ISUs 100 are aligned at the
point they reach the HDT 12. This is part of the upstream synchronization
process and shall be described further below. In addition, to maintain
orthogonality of the multicarriers, the carrier frequencies used for the
upstream transmission by the ISUs 100 must be frequency locked to the HDT
12.
Incoming downstream information from the CXMC 80 to the MCC modem 82 is
frame aligned to the 2 kHz and 8 kHz clocks provided to the MCC modem 82.
The 2 kHz multi-frame signal is used by the MCC modem 82 to convey
downstream symbol timing to the ISUs as described in further detail below.
This multiframe clock conveys the channel correspondence and indicates the
multi-carrier frame structure so that the telephony data may be correctly
reassembled at the ISU 100. Two kHz represents the greatest common factor
between 10 kHz (the modem symbol rate) and 8 kHz (the data frame rate).
All ISUs 100 will use the synchronization information inserted by the
associated MCC modem 82 to recover all downstream timing required by the
ISUs 100. This synchronization allows the ISUs 100 to demodulate the
downstream information and modulate the upstream transmission in such a
way that all ISU 100 transmissions received at the HDT 12 are synchronized
to the same reference. Thus, the carrier frequencies used for all ISU 100
upstream transmission will be frequency locked to the HDT 12.
The symbol alignment is performed over synchronization channels in the
downstream and upstream 6 MHz bandwidths under the responsibility of the
MCC modem 82, in addition to providing path delay adjustment,
initialization and activation, and provisioning over such synchronization
channels until initialization and activation is complete as further
described herein. These parameters are then tracked by use of the IOC
channels. Because of their importance in the system, the IOC channel and
synchronization channels may use a different modulation scheme for
transport of control data between the MCC modem 82 and ISUs 100 which is
more robust or of lesser order (less bits/sec/Hz or bits/symbol) than used
for transport of telephony data For example, the telephony data may be
modulated using quadrature amplitude modulation, while the IOC channel and
synchronization channel may be modulated utilizing BPSK modulation
techniques.
The MCC modem 82 also demodulates telephony and control data modulated on
multicarriers by the ISUs 100. Such demodulation is described further
below with respect to the various embodiments of the telephony transport
system.
Functions with respect to the OFDM transport system for which the MCC modem
82 is responsible, include at least the following, which are further
described with respect to the various embodiments in further detail. The
MCC modem 82 detects a received amplitude/level of a synchronization
pulse/pattern from an ISU 100 within a synchronization channel and passes
an indication of this level to the CXMC 80 over the communication
interface therebetween. The CXMC 80 then provides a command to the MCC
modem 82 for transmission to the ISU 100 being leveled for adjustment of
the amplitude level thereof. The MCC modem 82 also provides for symbol
alignment of all the upstream multicarriers by correlating an upstream
pattern modulated on a synchronization channel with respect to a known
symbol boundary and passing a required symbol delay correction to the CXMC
80 over the communication interface therebetween. The CXMC 80 then
transmits via the MCC modem 82 a message downstream to the ISU 100 to
adjust the symbol delay of the ISU 100.
Likewise, with regard to synchronizing an ISU 100 for overall path delay
adjustment, the MCC modem 82 correlates an upstream multiframe pattern
modulated in the proper bandwidth by the ISU 100 on the IOC channel with
respect to a known reference boundary, and passes a required path delay
correction to the CXMC 80 over the modem interface therebetween. The CXMC
80 then transmits via the MCC modem 82 over the IOC channel a message
downstream to adjust the overall path delay of an ISU 100.
Summary of Bidirectional Multi-Point to Point Telephony Transport
The following summarizes the transport of telephony and control information
over the HFC distribution network 11. Each CXMU 56 of HDT 12 is
provisioned with respect to its specific upstream and downstream operating
frequencies. The bandwidth of both upstream and downstream transmission by
the CXMU 56 are a maximum of 6 MHz, with the downstream transmission in a
6 MHz band of the RF spectrum of about 725-800 MHz
In the downstream direction, each MCC modem 82 of the CXMU 56 provides
electrical telephony and control data signals to the downstream telephony
transmitter 14 via coaxial line 22 in its provisional 6 MHz bandwidth. The
RF electrical telephony and control data signals from the MCC modems 82 of
the HDT 12 are combined into a composite signal. The downstream telephony
transmitter then passes the combined electrical signal to redundant
electrical-to-optical converters for modulation onto a pair of protected
downstream optical feeder lines 24.
The downstream optical feeder lines 24 carry the telephony information and
control data to an ODN 18. At the ODN 18, the optical signal is converted
back to electrical and combined with the downstream video information
(from the video head-end feeder line 42) into an electrical downstream RF
output signal. The electrical RF output signal including the telephony
information and control data is then fed to the four coaxial distribution
legs 30 by ODN 18. All telephony information and control data downstream
is broadcast on each coaxial leg 30 and carried over the coaxial portion
of the HFC distribution network 11. The electrical downstream output RF
signal is tapped from the coax and terminated on the receiver modem 101 of
an ISU 100 through diplex filter 104, shown in FIG. 8.
The RF electrical output signals include telephony information and control
data modulated on orthogonal multicarriers by MCC modem 82 utilizing
orthogonal frequency division multiplexing techniques; the telephony
information and control data being mapped into symbol data and the symbols
being modulated on a plurality of orthogonal carriers using fast Fourier
transform techniques. As the symbols are all modulated on carriers at a
single point to be transmitted to multiple points in the system 10,
orthogonality of the multicarriers and symbol alignment of the symbols
modulated on the orthogonal multicarriers are automatically aligned for
transport over the HFC distribution network 11 and the telephony
information and control data is demodulated at the ISUs 100 by the modem
101.
The ISU 100 receives the RF signal tapped from the coax of the coaxial
portion of the HFC network 11. The RF modem 101 of the ISU 100 demodulates
the signal and passes the telephony information and control data extracted
to the CXSU controller 102 for provision to channel units 103 as
appropriate. The ISU 100 represents the interface where the telephony
information is converted for use by a subscriber or customer.
The CXMUs 56 of the HDT 12 and the ISUs 100 implement the bidirectional
multi-point to point telephony transport system of the communication
system 10. The CXMUs 56 and the ISUs, therefore, carry out the modem
functionality. The transport system in accordance with the present
invention may utilize three different modems to implement the modem
functionality for the transport system. The first modem is the MCC modem
82 which is located in each CXMU 56 of the HDT 12. The HDT 12, for
example, includes three active MCC modems 82 (FIG. 3) and is capable of
supporting many ISUs 100, representing a multi-point to point transport
network. The MCC modem 82 coordinates telephony information transport as
well as control data transport for controlling the ISUs 100 by the HDT 12.
For example, the control data may include call processing messages,
dynamic allocation and assignment messages, ISU synchronization control
messages, ISU modem control messages, channel unit provisioning, and any
other ISU operation, administration, maintenance and provisioning (OAM&P)
information.
The second modem is a single family subscriber or HISU modem optimized to
support a single dwelling residential unit. Therefore, it must be low in
cost and low in power consumption. The third modem is the multiple
subscriber or MISU modem, which is required to generally support both
residential and business services.
The HISU modem and the MISU modem may take several forms. For example, the
HISU modem and the MISU modem may, as described further in detail below
with regard to the various embodiments of the present invention, extract
only a small portion of the multicarriers transmitted from the HDT 12 or a
larger portion of the multicarriers transmitted from the HDT 12. For
example, the HISU may extract 20 multicarriers or 10 payload channels of
telephony information transported from the HDT 12 and the MISU may extract
information from 260 multicarriers or 130 payload channels transported
from the HDT 12. Each of these modems may use a separate receiver portion
for extracting the control data from the signal transported by the HDT 12
and an additional receiver portion of the HISU modem to extract the
telephony information modulated on the multicarriers transported from the
HDT 12. This shall be referred to hereinafter as an out of band ISU modem.
The MCC modem 82 for use with an out of band ISU modem may modulate
control information within the orthogonal carrier waveform or on carriers
somewhat offset from such orthogonal carriers. In contrast to the out of
band ISU modem, the HISU and MISU modems may utilize a single receiver for
the ISU modem and extract both the telephony information and control data
utilizing the single receiver of the modem. This shall be referred to
hereinafter as an in-band ISU modem. In such a case, the control data is
modulated on carriers within the orthogonal carrier waveform but may
utilize different carrier modulation techniques. For example, BPSK for
modulation of control data on the carriers as opposed to modulation of
telephony data on payload carriers by QAM techniques. In addition,
different modulation techniques may be used for upstream or downstream
transmission for both control data and telephony data For example,
downstream telephony data may be modulated on the carriers utilizing 256
QAM and upstream telephony data may be modulated on the carriers utilizing
32 QAM. Whatever modulation technique is utilized for transmission
dictates what demodulation approach would be used at the receiving end of
the transport system. Demodulation of the downstream telephony information
and control data transported by the HDT 12 shall be explained in further
detail below with reference to block diagrams of different modem
embodiments.
In the upstream direction, each ISU modem 101 at an ISU 100 transmits
upstream on at least one orthogonal multicarrier in a 6 MHz bandwidth in
the RF spectrum of about 5 to 40 MHz; the upstream 6 MHz band
corresponding to the downstream 6 MHz band in which transmissions are
received. The upstream electrical telephony and control data signals are
transported by the ISU modems 101 to the respectively connected optical
distribution node 18 as shown in FIG. 1 via the individual coaxial cable
legs 30. At the ODN 18, the upstream signals from the various ISUs are
combined and transmitted optically to the HDT 12 via optical feeder lines
26. As previously discussed, the upstream electrical signals from the
various ISUs may, in part, be frequency shifted prior to being combined
into a composite upstream optical signal. In such a case, the telephony
receiver 16 would include corresponding downshifting circuitry.
Due to the multi-point to point nature of transport over the HFC
distribution network 11 from multiple ISUs 100 to a single HDT 12, in
order to utilize orthogonal frequency division multiplexing techniques,
symbols modulated on each carrier by the ISUs 100 must be aligned within a
certain phase margin. In addition, as discussed in further detail below,
the round trip path delay from the network interface 62 of the HDT 12 to
all ISUs 100 and back from the ISUs 100 to the network interface 62 in the
communication system 10 must be equal. This is required so that signaling
multiframe integrity is preserved throughout the system. In addition, a
signal of proper amplitude must be received at the HDT 12 to perform any
control functions with respect to the ISU 100. Likewise, with regard to
OFDM transport from the ISUs 100, the ISUs 100 must be frequency locked to
the HDT 12 such that the multicarriers transported over the HFC
distribution network 11 are orthogonally aligned. The transport system
implements a distributed loop technique for implementing this multi-point
to point transport utilizing orthogonal frequency division multiplexing as
further described below. When the HDT 12 receives the plurality of
multicarriers which are orthogonally aligned and which have telephony and
control data modulated thereon with symbols aligned, the MCC modems 82 of
the CXMUs 56 demodulate the telephony information and control data from
the plurality of multicarriers in their corresponding 6 MHz bandwidth and
provide such telephony data to the CTSU 54 for delivery to the network
interface 62 and the control data to the CXMC 80 for control of the
telephony transport.
As one skilled in the art will recognize, the spectrum allocations,
frequency assignments, data rates, channel numbers, types of services
provided and any other parameters or characteristics of the system which
may be a choice of design are to be taken as examples only. The invention
as described in the accompanying claims contemplates such design choices
and they therefore fall within the scope of such claims. In addition, many
functions may be implemented by software or hardware and either
implementation is contemplated in accordance with the scope of the claims
even though reference may only be made to implementation by one or the
other.
First Embodiment of Telephony Transport System
The first embodiment of the telephony transport system in accordance with
the present invention shall be described with particular reference to
FIGS. 13-35 which include block diagrams of MCC modems 82, and HISU modems
and MISU modems shown generally as ISU modem 101 in FIG. 8. Such modems
implement the upstream and downstream modem transport functionality.
Following this description is a discussion on the theory of operation
utilizing such modems.
Referring to FIG. 13, the spectrum allocation for one 6 MHz band for
upstream and downstream transport of telephony information and control
data utilizing OFDM techniques is shown. The waveform preferably has 240
payload channels or DS0+ channels which include 480 carriers or tones for
accommodating a net data rate of 19.2 Mbps, 24 IOC channels including 46
carriers or tones, and 2 synchronization channels. Each synchronization
channel includes two carriers or tones and is each offset from 24 IOC
channels and 240 payload channels by 10 unused carriers or tones, utilized
as guard tones. The total carriers or tones is 552. The synchronization
tones utilized for synchronization functions as described further below
are located at the ends of the 6 MHz spectrum and the plurality of
orthogonal carriers in the 6 MHz band are separated from carriers of
adjacent 6 MHz bands by guard bands (516.0 kHz) at each end of the 6 MHz
spectrum. The guard bands are provided at each end of the 6 MHz band to
allow for filter selectivity at the transmitter and receivers of the
system. The synchronization carriers are offset from the telephony data or
payload carriers such that if the synchronization carrier utilized for
synchronization during initialization and activation is not orthogonal
with the other tones or carriers within the 6 MHz band, the
synchronization signal is prevented from destroying the structure of the
orthogonally aligned waveform. The synchronization tones are, therefore,
outside of the main body of payload carriers of the band and interspersed
IOC channels, although the synchronization channel could be considered a
special IOC channel.
To minimize the power requirement of the ISUs, the amount of bandwidth that
an ISU processes is minimized. As such, the telephony payload channels and
IOC channels of the 6 MHz band are interspersed in the telephony payload
channels with an IOC channel located every 10 payload channels. With such
a distributed technique, wherein subbands of payload channels greater than
10 include an IOC channel, the amount of bandwidth an ISU "sees" can be
limited such that an IOC channel is available for the HDT 12 to
communicate with the ISU 100. Such subband distribution for the spectral
allocation shown in FIG. 13 is shown in FIG. 16. There are 24 subbands in
the 6 MHz bandwidth with each subband including 10 payload channels with
an IOC channel between the 5th and 6th payload channels. A benefit of
distributing the IOC channels throughout the 6 MHz band is protection from
narrow band ingress. If ingress destroys an IOC channel, there are other
IOC channels available and the HDT 12 can re-tune an ISU 100 to a
different portion of the 6 MHz band, where an IOC channel that is not
corrupted is located.
Preferably, the MISU 66 sees approximately 3 MHz of the 6 MHz bandwidth to
receive up to 130 payload channels which bandwidth also includes numerous
IOC channels for communication from the HDT 12 to the MISU 66. The HISU 68
sees about 100 kHz of the 6 MHz bandwidth to receive 11 channels including
at least one IOC channel for communication with the HDT 12.
The primary difference between the downstream and upstream paths are the
support of downstream synchronization and upstream synchronization. In the
downstream direction, all ISUs lock to information from the HDT (point to
multi-point). The initialization and activation of ISUs are based on
signals supplied in the upstream synchronization channel. During
operation, ISUs track the synchronization via the IOC channels. In the
upstream, the upstream synchronization process involves the distributed
(multi-point to point) control of amplitude, frequency, and timing;
although frequency control can also be provided utilizing only the
downstream synchronization channel as described further below. The process
of upstream synchronization occurs in one of the two upstream
synchronization channels, the primary or the secondary synchronization
channel.
Referring to FIG. 21, the downstream transmission architecture of the MCC
modem 82 is shown. Two serial data inputs, approximately 10 Mbps each,
comprise the payload data from the CXMC 56 which is clocked by the 8 kHz
frame clock input. The IOC control data input from the CXMC 56 is clocked
by the IOC clock input, which is preferably a 2.0 kHz clock. The telephony
payload data and the IOC control data enter through serial ports 132 and
the data is scrambled as known to one skilled in the art by scrambler 134
to provide randomness in the waveform to be transmitted over the HFC
distribution network 11. Without scrambling, very high peaks in the
waveform may occur, however, if the waveform is scrambled the symbols
generated by the MCC modem 82 become sufficiently random and such peaks
are sufficiently limited.
FIG. 58 details the operation of a typical scrambler, such as 134, FIG. 21.
Symbol clock 4501 clocks a seed pattern through a linear-feedback shift
register 4510 having nine stages, 4510-0 through 4510-8. With XOR gate
4511 positioned as shown, the generator polynomial is binary "100 010
000". The seed initially loaded into register 4510 at input 4502 is "111
001 110". Two identical translation tables 4520 and 4521 receive two-bit
inputs from register 4510 at every symbol time. The high- and low-order
bits of table 4520 proceed from the outputs of stages 4510-7 and 4510-6,
respectively. High-order bit 4523 of table 4521 also receives output
4510-6, but as its high-order bit; stage output 4510-5 provides its
low-order bit. Logic gates 4530 perform an XOR between the five-bit output
of table 4520 and the upper five bits of a 10-bit DS0 word, while gates
4531 do the same for the lower five bits of the same DS0 word. Outputs
4505 and 4506 carry the two 5-bit scrambled symbols for the DS0 word. Each
descrambler such as 176, FIG. 22 or 23, is identical to its corresponding
scrambler. It recovers the original bit pattern of each symbol by decoding
it with the same polynomial and seed.
The polynomial and seed for register 4510 of the scramblers and
descramblers selected by known techniques to yield a maximal-length
pseudo-random sequence. Inversion of the order of the input bits as
between table 4520 and table 4521 increases the scrambling of the two
symbols of the DS0 word. To increase the randomness among different
sequences even more, different scramblers in the system have different
polynomials and seeds. Randomness could be further increased by using more
than four different table entries; however, the added complexity overrode
the gain, for this particular embodiment Only the payload channels are
scrambled; the IOC channels are not scrambled.
The scrambled signals are applied to a symbol mapping function 136. The
symbol mapping function 136 takes the input bits and maps them into a
complex constellation point. For example, if the input bits are mapped
into a symbol for output of a BPSK signal, every bit would be mapped to a
single symbol in the constellation as in the mapping diagram for BPSK of
FIG. 15. Such mapping results in in-phase and quadrature values (I/Q
values) for the data BPSK is the modulation technique preferably used for
the upstream and downstream IOC channels and the synchronization channels.
BPSK encoding is preferred for the IOC control data so as to provide
robustness in the system as previously discussed. For QPSK modulation,
every two bits would map into one of four complex values that represent a
constellation point. In the preferred embodiment, 32 QAM is utilized for
telephony payload data, wherein every five bits of payload data is mapped
into one of 32 constellation points as shown in FIG. 14. Such mapping also
results in I/Q values. As such, one DS0+ signal (10 bits) is represented
by two symbols and the two symbols are transmitted using two carriers.
Thus, one DS0+ channel is transported over two carriers or tones of 6 MHz
spectrum.
One skilled in the art will recognize that various mapping or encoding
techniques may be utilized with different carriers. For example, telephony
channels carrying ISDN may be encoded using QPSK as opposed to telephony
channels carrying POTS data being encoded using 32 QAM. Therefore,
different telephony channels carrying different services may be modulated
differently to provide for more robust telephony channels for those
services that require such quality. The architecture in accordance with
the present invention provides the flexibility to encode and modulate any
of the channels differently from the modulation technique used for a
different channel.
Within the framework of QAM32 modulation, FIG. 17 shows a constellation
which has improved characteristics. Here, the in-phase and quadrature
values are shown encoded by three bits each instead of the four shown in
FIG. 13; their analog values, however, they remain in the ranges -5 to +5.
The constellation of FIG. 17 approaches as closely as possible to an
analogy to a Gray code scheme, in which a transition from one row to the
next and from one column to the next result in only a single bit change in
the 5-bit symbol code. (The exceptions are four transitions from the first
column to the second, and from the fifth to the sixth, which have two
transitions each. The corner cells have zero transitions between these
columns, which do not detract from the advantages of the scheme.) If a
symbol is received incorrectly after transmission, the most likely error
is a slight change in either amplitude or phase. If the bit strings
represented by the symbols have as few bit transitions as possible for
single-value phase and amplitude changes, then a reception error will
create fewer bit errors on the final digital output. That is, small
(symbol) errors in produce small (bit) errors out.
The constellations shown in FIGS. 14 and 17 use all points of a six-cell
square except the four corners. Hence, they have two axes of symmetry, and
appear identical when rotated by 90.degree., 180.degree., and 270.degree..
If a phase error ever exceeds 45.degree., an attempted correction may pull
the phase to an incorrect orientation. This is called four-fold phase
ambiguity. However, deliberately using one and only one of the corner
points as a valid symbol provides a key for identifying the correct phase
for errors as great as a full 180.degree.. For example, designating the
symbol for "16" as I=010 (+5) and Q=010 (+5) instead of the I=001, Q=010
(+3, +5) in FIG. 17 introduces a symbol at this corner point whenever a
"16" is sent upstream or downstream. Because only one corner is used, any
received value having both I and Q values .+-.5 requires phase rotation
until I=+5 and Q=+5. This assignment also preserves the nearly Gray-code
structure of the constellation.
Any other symbol assignment which breaks the symmetry of the constellation
would produce the same effect. Even a constellation retaining only one
axis of symmetry would allow twice the phase-correction range of the
constellation of FIG. 17. For example, using both the upper left and lower
right corners as valid symbols allows correction of phase errors up to
90.degree..
Each symbol that gets represented by the I/Q values is mapped into a fast
Fourier transform (FFT) bin of symbol buffer 138. For example, for a DS0+,
running at 8 kHz frame rate, five bits are mapped into one FFT bin and
five bits into another bin Each bin or memory location of the symbol
buffer 138 represents the payload data and control data in the frequency
domain as I/Q values. One set of FFT bins gets mapped into the time domain
through the inverse FFT 140, as is known to one skilled in the art The
inverse FFT 140 maps the complex I/Q values into time domain samples
corresponding to the number of points in the FFT. Both the payload data
and IOC data are mapped into the buffer 138 and transformed into time
domain samples by the inverse FFT 140. The number of points in the inverse
FFT 140 may vary, but in the preferred embodiment the number of points is
256. The output of the inverse FFT 140, for a 256 point FFT, is 256 time
domain samples of the waveform.
In conventional practice, buffer 138 clocks symbols into inverse FFT 140 at
exactly the same rate that inverse FFT 140 clocks out the in-phase and
quadrature values FFT I and FFT Q in FIG. 21. To put the matter another
way, the 256 digital waveform samples from buffer 138 represent
360.degree., or 2.pi. radians, of a QAM 32 waveform having the amplitude
and phase of the 5 bits of its symbol, as determined by mapping unit 136.
The FFT I and Q outputs represent 256 samples of a frequency spectrum
corresponding to the same time period. At the receiving end, however, any
misalignment at all in the phase synchronization causes FFT 170, FIG. 22,
or 180, FIG. 23, to decode a portion of a previous or subsequent symbol's
waveform along with somewhat less than the full cycle of the desired
symbol; this intersymbol interference can cause misreading the symbol as a
different valid symbol, resulting in as many as five bit errors.
In a presently preferred embodiment, the 256 samples clocked into inverse
FFT 140 represent an extra 45.degree. (.pi./4 radians) above a complete
cycle. Another way to think of this is that the symbols are clocked into
the FFT at an effective 9 kHz rate, and clocked out at the nominal 8 kHz
symbol rate. FIG. 52 shows an unmodulated sine wave, i.e., one having 1=0,
Q=0 in the units used herein. The upper portion shows one cycle,
0-360.degree., at the nominal 8 kHz frame rate. The lower portion shows
the same wave at a 9 kHz rate, so that the amount of time previously
occupied by 360.degree. now takes up 405.degree. of phase--from
-22.5.degree. to +382.5.degree.. Obviously, there are phase
discontinuities between successive cycles of the wave. FIG. 53 shows a
typical QAM 32 wave modulated at a different amplitude and a slightly
different phase from those of FIG. 52. These might correspond to, say,
I=1, Q=+1 in the scheme used herein. The small portions at the ends of
this wave represent unmodulated cycles, as in FIG. 52. The phase of this
wave is advanced from the corresponding wave of the lower portion of FIG.
52; it does not cross the zero axis at 0.degree. and 180.degree. of its
proper cycle. It does, however, include the extra 22.5.degree. of excess
phase at each end, for 45.degree. extra over an 8 kHz cycle. Again, a
phase discontinuities exist at the ends of the total 405.degree. phase
degrees of this wave.
In fact, this characteristic gives the excess-phase improvement an
advantage over its primary function of providing a guard band for the
symbol decoder, for reducing intersymbol interference. In FIG. 52,
successive cycles of a wave modulated with the same symbol (or with no
symbol), produce a continuous waveform with no breaks or other features to
distinguish the beginnings and endings of individual cycles. The lower
part of this diagram demonstrates that even an unmodulated excess-phase
waveform contains discontinuity features serving as markers at the ends of
each cycle. A repeating string of idle symbols, or any other symbols,
likewise produces these markers. In the frequency- and phase-acquisition
and tracking aspects discussed below, such markers therefore provide
definite waveform features for synchronizing purposes, without having to
guarantee the transmission of any special string of varying characters
strictly for synchronization. This saves the overhead of interrupting the
payload and/or IOC channels to provide such a string, and the complexity
of storing or diverting payload information while the sync string is
present. It also allows sync to take place at times when, because of the
above factors, it would not be feasible otherwise.
At the receiving end, FFT 170 (in an MISU) or 180 (HISU) decodes the 256
time slots for one frame time as 405.degree. of a cycle to symbol decoder
174, which matches the cycle to the nearest 5-bit string of bits. Because
any phase difference up to .+-.22.5.degree. will never conflate the proper
wave with that for another symbol, no intersymbol interference at all
occurs within this margin of error in phase tracking. This provides a form
of guard band for each symbol. In the upstream direction, units 186, 188,
and 190 or 191 provide excess phase in the transmitting MISU and HISU
modems of FIGS. 24 and 25; and the head-end receiving modem of FIG. 26
decodes and tracks this phase as described above.
The inverse FFT 140 has separate serial outputs for in-phase and quadrature
(I/Q) components, FFT1 and FFT0. Digital to analog converters 142 take the
in-phase and quadrature components, which is a numeric representation of
baseband modulated signal and convert it to a discrete waveform. The
signal then passes through reconstruction filters 144 to remove harmonic
content. This reconstruction is needed to avoid problems arising from
multiple mixing schemes and other filtering problems. The signal is summed
in a signal conversion transmitter 146 for up-converting the I/Q
components utilizing a synthesized waveform that is digitally tunable with
the in-phase and quadrature components for mixing to the applicable
transmit frequency. For example, if the synthesizer is at 600 MHz, the
output frequency will be at 600 MHz. The components are summed by the
signal conversion transmitter 146 and the waveform including a plurality
of orthogonal carriers is then amplified by transmitter amplifier 148 and
filtered by transmitter filter 150 before being coupled onto the optical
fiber by way of telephony transmitter 14. Such functions are performed
under control of general purpose processor 149 and other processing
circuitry of block 147 necessary to perform such modulation. The general
purpose processor also receives ISU adjustment parameters from carrier,
amplitude, timing recovery block 222 (FIG. 26) for carrying out
distributed loop symbol alignment, frequency locking, amplitude
adjustment, and path delay functions as described further below.
In conventional practice, the relationship between the frequency of a
carrier and the frequency and timing of data symbols modulated onto that
carrier is arbitrary and unimportant In the present system, however, it
has been found that even very small frequency drifts between the 8 kHz
symbol or frame clock and the frequencies of the tones upon which they
ride can produce significant intersymbol interference and distortion at
the receiving end. Such drifts tend to destroy the orthogonality of the
channel signals produced by inverse FFT 140 in FIG. 21. The present system
also, however, provides a simple, inexpensive way to overcome this
problem. FIG. 51 shows a portion 4200 of the HDT clock/sync logic in CTSU
54, FIG. 3. Timing recovery loop 4210 produces a single master reference
clock output at 10.24 MHz Although loop 4210 could be a free-running
oscillator, it is in fact slaved to the network 10, FIG. 1. With which the
entire system communicates. This connection is convenient in eliminating
gross or unpredictable differences between the data speeds of the network
and the system.
Smoothing loop 4220 evens out short-term variations in the signal from loop
4210. Phase comparator 4221 controls a voltage-controlled crystal
oscillator at 40.96 MHz; divider 4223 provides feedback at the proper
frequency. Comparator 4221 includes a low-pass integrator which gives
phase-lock loop 4220 a bandwidth of about 130 Hz Divider 4230 reduces the
frequency of VCXO 4222 to 2.56 Mhz A second phase-lock loop 4240 has a
phase comparator 4241, again with low-pass characteristics, feeding a
voltage-controlled oscillator running at 1267.2 MHz; divider provides
feedback at the proper frequency. Divider 4250 produces the final RF clock
frequency, 9.9 MHz at output 4251. The network clock is sufficiently
accurate over long periods of time, but it is subject to significant
amounts of short-period jitter. The large amount of smoothing provided by
loops 4220 and 4240 overcome the intolerance of analog RF components for
short-term variations.
Meanwhile, digital divider 4260 divides the master 10.24 MHz clock by a
factor of 80 to produce an 8 kHz symbol or frame clock output 4261. Output
4261 does not require the smoothing, because it clocks only digital
circuits, which are relatively insensitive to short-term frequency
changes.
RF master clock 4261 proceeds to RF synthesizer 143 in HDT transmitting
modem 82, as shown in FIG. 21, where it directly controls the frequency of
the tunable 500-850 MHz RF carrier for the entire band carrying all of the
channels shown in FIGS. 13 and 16. Symbol clock 4261 proceeds to the
frame-clock inputs in FIG. 21, where it controls the symbol timing, and,
because it also controls the FFT speed, the frequencies of the channels in
the entire band. Clock lock 4200 thus provides a solid link which
inherently preserves the orthogonality of the band signals in a
multicarrier system, by deriving the RF carrier clock and the symbol or
frame clock from the same source. At the same time, it provides a small
amount of gradual variation for satisfying the demands of the analog RF
components.
The overall purpose of locking the two clocks together at the HDT is to
lock the carrier clocks and the symbol (frame) clocks throughout the
system; and the purpose of this in turn is to preserve the orthogonality
of the signals in a multicarrier system which is capable of bidirectional
operation: that is, as a multipoint-to-point-configuration as well as in
the usual point-to-multipoint "broadcast" direction. Clock generator 166,
FIGS. 22 and 23, of timing generator 107, FIG. 6 locks to the frequencies
of the incoming signals to provide the clocks used in the remote ISU
modules. Therefore, the carrier and frame clocks in each upstream
transmitter portion, FIG. 24, of remote modem 108, FIG. 8, are also locked
to each other, by virtue of being locked to the incoming signal from the
HDT.
At the downstream receiving end, either an MISU or an HISU provides for
extracting telephony information and control data from the downstream
transmission in one of the 6 MHz bandwidths. With respect to the MISU 66,
the MISU downstream receiver architecture is shown in FIG. 22. It includes
a 100 MHz bandpass filter 152 to reduce the frequency band of the received
600 to 850 MHz total band broadcast downstream. The filtered signal then
passes through voltage tuned filters 154 to remove out of band
interference and further reduce the bandwidth. The signal is down
converted to baseband frequency via quadrature and in-phase down converter
158 where the signal is mixed at complex mixers 156 utilizing synthesizer
157 which is controlled from an output of serial ports 178. The down
converted I/Q components are passed through filters 159 and converted to
digital format at analog to digital converters 160. The time domain
samples of the I/Q components are placed in a sample buffer 162 and a set
of samples are input to down converter compensation unit 164. The
compensation unit 164 attempts to mitigate errors such as DC offsets from
the mixers and differential phase delays that occur in the down
conversion.
Carrier, amplitude and timing signaling are exacted from the compensated
signal, by the carrier, amplitude, and timing recovery block 166 by
extracting control data from the synchronization channels during
initialization and activation of the ISU and the IOC channels during
tracking as further described below with reference to FIG. 33. The
compensated signal in parallel form is provided to fast Fourier transform
(FFT) 170 to be converted into a vector of frequency domain elements which
are essentially the complex constellation points with I/Q components
originally created upstream at the MCC modem 82 for the DS0+ channels
which the MISU sees. Due to inaccuracies in channel filtering, an
equalizer 172 removes dynamic errors that occur during transmission and
reception. Equalization in the upstream receiver and the downstream
receiver architectures shall be explained in further detail below with
reference to FIG. 35. From the equalizer 172, the complex constellation
points are converted to bits by symbol to bit converter 174, descrambled
at descrambler 176 which is a mirror element of scrambler 134, and the
payload telephony information and IOC control data are output by the
serial ports 178 to the CXSU 102 as shown in FIG. 8. Block 153 includes
the processing capabilities for carrying out the various functions as
shown therein.
Referring to FIG. 23, the HISU 68 downstream receiver architecture is
shown. The primary difference between the HISU downstream receiver
architecture (FIG. 23) and the MISU downstream receiver architecture (FIG.
22) is the amount of bandwidth being processed. The front ends of the
receivers up to the FFT processing are substantially the same, except
during the down conversion, the analog to digital converters 160 can be
operated at a much slower rate. For instance, if the bandwidth of the
signal being processed is 100 kHz, the sample rate can be approximately
200 kHz. In an MISU processing a 3 MHz signal the sample rate is about 6
MHz. Since the HISU is limited to receiving a maximum of 10 DS0+s, the FFT
180 can be of a smaller size. A 32 point FFT 180 is preferably used in the
HISU and can be implemented more efficiently, compared to a 128 or 256
point FFT utilized in the MISU. Therefore, the major difference between
these architectures is that the HISU receiver architecture requires
substantially less signal processing capability than the MISU receiver and
as such has less power consumption Thus, to provide a system wherein power
consumption at the remote units is minimized, the smaller band of
frequencies seen by the HISU allows for such low consumption. One reason
the HISU is allowed to see such a small band of carriers is that the IOC
channels are interspersed throughout the 6 MHz spectrum.
Referring to FIG. 24, the upstream transmission architecture for the HISU
68 is shown. The IOC control data and the telephony payload data from the
CXSU 102 (FIG. 8) is provided to serial ports 182 at a much slower rate in
the HISU than in the MISU or HDT transmission architectures, because the
HISU supports only 10 DS0+ channels. The HISU upstream transmission
architecture implements three important operations. It adjusts the
amplitude of the signal transmitted, the timing delay (both symbol and
path delay) of the signal transmitted, and the carrier frequency of the
signal transmitted. The telephony data and IOC control data enters through
the serial ports 182 under control of clocking signals generated by the
clock generator 173 of the HISU downstream receiver architecture, and is
scrambled by scrambler 184 for the reasons stated above with regard to the
MCC downstream transmission architecture. The incoming bits are mapped
into symbols, or complex constellation points, including I/Q components in
the frequency domain, by bits to symbol converter 186. The constellation
points are then placed in symbol buffer 188. Following the buffer 188, an
inverse FFT 190 is applied to the symbols to create time domain samples;
32 samples corresponding to the 32 point FFT. A delay buffer 192 is placed
on the output of the inverse FFT 190 to provide multi-frame alignment at
MCC modem upstream receiver architecture as a function of the upstream
synchronization process controlled by the HDT 12. The delay buffer 192,
therefore, provides a path delay adjustment prior to digital to analog
conversion by the digital to analog converters 194 of the in-phase and
quadrature components of the output of the inverse FFT 190. Clock delay
196 provides a fine tune adjustment for the symbol alignment at the
request of IOC control data output obtained by extracting control data
from the serial stream of data prior to being scrambled. After conversion
to analog components by digital to analog converters 194, the analog
components therefrom are reconstructed into a smooth analog waveform by
the reconstruction filters 198. The upstream signal is then directly up
converted by direct converter 197 to the appropriate transmit frequency
under control of synthesizer block 195. Synthesizer block 195 is operated
under control of commands from an IOC control channel which provides
carrier frequency adjustment commands thereto as extracted in the HISU
downstream receiver architecture. The up converted signal is then
amplified by transmitter amplifier 200, filtered by transmitter filter 202
and transmitted upstream to be combined with other signals transmitted by
other ISUs 100. The block 181 includes processing circuitry for carrying
out the functions thereof.
Referring to FIG. 27, the upstream transmitter architecture for the MISU 66
is shown and is substantially the same as the upstream transmitter
architecture of HISU 68. However, the MISU 66 handles more channels and
cannot perform the operation on a single processor as can the HISU 68.
Therefore, both a processor of block 181 providing the functions of block
181 including the inverse FFT 190 and a general purpose processor 206 to
support the architecture are needed to handle the increased channel
capacity.
Referring to FIG. 26, the MCC upstream receiver architecture of each CXMU
56 at the HDT 12 is shown. A 5 to 40 MHz band pass filter 208 filters the
upstream signal which is then subjected to a direct down conversion to
baseband by mixer and synthesizer circuitry 211. The outputs of the down
conversion is applied to anti-alias filters 210 for conditioning thereof
and the output signal is converted to digital format by analog to digital
converters 212 to provide a time domain sampling of the in-phase and
quadrature components of the signal to narrow band ingress filter and FFT
112. The narrow band ingress filter and FFT 112, as described below,
provides protection against narrow band interference that may affect the
upstream transmission.
The ingress filter and FFT 112 protects ten channels at a time, therefore,
if ingress affects one of the available 240 DS0+s in the 6 MHz spectrum
received by MCC modem 82, a maximum of ten channels will be destroyed from
the ingress. The ingress filter and FFT 112 includes a polyphase
structure, as will be recognized by one skilled in the art as a common
filter technique. It will be further recognized by one skilled in the art
that the number of channels protected by the polyphase filter can be
varied. The output of the ingress filter and FFT 112 is coupled to an
equalizer 214 which provides correction for inaccuracies that occur in the
channel, such as those due to noise from reference oscillators or
synthesizers. The output symbols of the equalizer 214, are applied to a
symbols to bits converter 216 where the symbols are mapped into bits. The
bits are provided to descramblers 218, which are a mirror of the
scramblers of the ISUs 100 and the output of the descramblers are provided
to serial ports 220. The output of the serial ports is broken into two
payload streams and one IOC control data stream just as is provided to the
MCC downstream transmitter architecture in the downstream direction. Block
217 includes the necessary processing circuitry for carrying out the
functions therein.
In order to detect the downstream information, the amplitude, frequency,
and timing of the arriving signal must be acquired using the downstream
synchronization process. Since the downstream signal constitutes a point
to multi-point node topology, the OFDM waveform arrives via a single path
in an inherently synchronous manner, in contrast to the upstream signal.
Acquisition of the waveform parameters is initially performed on the
downstream synchronization channels in the downstream synchronization
bands located at the ends of the 6 MHz spectrum. These synchronization
bands include a single synchronization carrier or tone which is BPSK
modulated by a 2 kHz framing clock This tone is used to derive initial
amplitude, frequency, and timing at the ISU. The synchronization carrier
may be located in the center of the receive band and could be considered a
special case of an IOC. After the signal is received and the receiver
architecture is tuned to a typical IOC channel, the same circuitry is used
to track the synchronization parameters using the IOC channel. The process
used to acquire the necessary signal parameters utilizes carrier,
amplitude and timing recovery block 166 of the ISU receiver architecture,
which is shown in more detail in block diagram form in FIG. 33. The
carrier, amplitude and timing recovery block 166 includes a Costas loop
330 which is used to acquire the frequency lock for the received waveform.
After the signal is received from the compensation unit 164, a sample and
hold 334 and analog to digital conversion 332 is applied to the signal
with the resulting samples from the converters 332 applied to the Costas
loop 330. The sampling is performed under control of voltage controlled
oscillator 340 as divided by divider 333 which divides by the number of
points of the FFT utilized in the receiver architecture, M. The mixers 331
of the Costas loop 330 are fed by the arriving signal and the feedback
path, and serve as the loop phase detectors. The output of the mixers 331
are filtered and decimated to reduce the processing requirements of
subsequent hardware. Given that the received signal is band-limited, less
samples are required to represent the synchronization signal. If
orthogonality is not preserved in the receiver, the filter will eliminate
undesired signal components from the recovery process. Under conditions of
orthogonality, the LPF 337 will completely remove effects from adjacent
OFDM carriers. When carrier frequency lock is achieved, the process will
reveal the desired BPSK waveform in the in-phase arm of the loop. The
output of the decimators are fed through another mixer, then processed
through the loop filter with filter function H(s) and numerically
controlled oscillator (NCO), completing the feedback path to correct for
frequency error. When the error is at a "small" level, the loop is locked.
In order to achieve fast acquisition and minimal jitter during tracking,
it will be necessary to employ dual loop bandwidths. System operation will
require that frequency lock is achieved and maintained within about .+-.4%
of the OFDM channel spacing (360 Hz).
The amplitude of the signal is measured at the output of the frequency
recovery loop at BPSK power detector 336. The total signal power will be
measured, and can be used to adjust a numerically controllable analog gain
circuit (not shown). The gain circuit is intended to normalize the signal
so that the analog to digital converters are used in an optimal operating
region.
Timing recovery is performed using an early-late gate type algorithm of
early-late gate phase detector 338 to derive timing error, and by
adjusting the sample clock or oscillator 340 in response to the error
signal. The early-late gate detector results in an advance/retard command
during an update interval. This command will be applied to the sample
clock or oscillator 340 through filter 341. This loop is held off until
frequency lock and amplitude lock have been achieved. When the timing loop
is locked, it generates a lock indicator signal. The same clocks are also
used for the upstream transmission. The carrier, timing and amplitude
recovery block 166 provides a reference for the clock generator 168. The
clock generator 168 provides all of the clocks needed by the MISU, for
example, the 8 kHz frame clock and the sample clock.
Carrier, amplitude, and timing recovery block 222 of the MCC modem upstream
receiver architecture (FIG. 26), is shown by the synchronization loop
diagram of FIG. 34. It performs detection for upstream synchronization on
signals on the upstream synchronization channel. For initialization and
activation of an ISU, upstream synchronization is performed by the HDT
commanding one of the ISUs via the downstream IOC control channels to send
a reference signal upstream on a synchronization channel. The carrier,
amplitude, and timing recovery block 222 measures the parameters of data
from the ISU 100 that responds on the synchronization channel and
estimates the frequency error, the amplitude error, and the timing error
compared to references at the HDT 12. The output of the carrier,
amplitude, and timing recovery block 222 is turned into adjustment
commands by the HDT 12 and sent to the ISU being initialized and activated
in the downstream direction on an IOC control channel by the MCC
downstream transmitter architecture.
The purpose of the upstream synchronization process is to initialize and
activate ISUs such that the waveform from distinct ISUs combine to a
unified waveform at the HDT 12. The parameters that are estimated at the
HDT 12 by carrier, amplitude, and timing recovery block 222 and adjusted
by the ISUs are amplitude, timing, and frequency. The amplitude of an ISUs
signal is normalized so that DS0+s are apportioned an equal amount of
power, and achieves a desired signal to noise ratio at the HDT 12. In
addition, adjacent ISUs must be received at the correct relative level or
else weaker DS0+ channels will be adversely impacted by the transient
behavior of the stronger DS0+ channels. If a payload channel is
transmitted adjacent to another payload channel with sufficient frequency
error, orthogonality in the OFDM waveform deteriorates and error rate
performance is compromised. Therefore, the frequency of the ISU must be
adjusted to close tolerances. Timing of the recovered signal also impacts
orthogonality. A symbol which is not aligned in time with adjacent symbols
can produce transitions within the part of the symbol that is subjected to
the FFT process. If the transitions of all symbols don't fall within the
guard interval at the HDT, approximately .+-.16 tones (8 DS0+s) relative
to the non-orthogonal channel will be unrecoverable.
During upstream synchronization, the ISUs will be commanded to send a
signal, for example a square wave signal, to establish amplitude and
frequency accuracy and to align symbols. The pattern signal may be any
signal which allows for detection of the parameters by carrier, amplitude
and timing recovery block 222 and such signal may be different for
detecting different parameters. For example, the signal may be a
continuous sinusoid for amplitude and frequency detection and correction
and a square wave for symbol timing. The carrier, amplitude and timing
recovery block 222 estimates the three distributed loop parameters. In all
three loops, the resulting error signal will be converted to a command by
the CXMC 80 and sent via the MCC modem 82 over an IOC channel and the CXSU
will receive the command and control the adjustment made by the ISU.
As shown in FIG. 34, the upstream synchronization from the ISU is sampled
and held 434 and analog to digital converted 432 under control of voltage
controlled oscillator 440. Voltage controlled oscillator is a local
reference oscillator which is divided by M, the points of the FFT in the
receiver architecture, for control of sample and hold 434 and analog to
digital converter 432 and divided by k to apply an 8 kHz signal to phase
detector 438.
Frequency error may be estimated utilizing the Costas loop 430. The Costas
loop 430 attempts to establish phase lock with the locally generated
frequency reference. After some period of time, loop adaptation will be
disabled and phase difference with respect to the time will be used to
estimate the frequency error. The frequency error is generated by filter
function H(s) 444 and provided to the CXMC 80 for processing to send a
frequency adjustment command to the ISU via an IOC control channel. The
frequency error is also applied to the numerically controlled oscillator
(NCO) to complete the frequency loop to correct for frequency error.
The amplitude error is computed based on the magnitude of the carrier
during the upstream synchronization by detecting the carrier amplitude of
the in-phase arm of the Costas loop 430 by power detector 436. The
amplitude is compared with a desired reference value at reference
comparator 443 and the error will be sent to the CXMC 80 for processing to
send an amplitude adjustment command to the ISU via an IOC control
channel.
When the local reference in the HDT has achieved phase lock, the BPSK
signal on the synchronization channel arriving from the ISU is available
for processing. The square wave is obtained on the in-phase arm of the
Costas loop 430 and applied to early-late gate phase detector 438 for
comparison to the locally generated 8 kHz signal from divider 435. The
phase detector 438 generates a phase or symbol timing error applied to
loop filter 441 and output via line 439. The phase or symbol timing error
is then provided to the CXMC 80 for processing to send a symbol timing
adjustment command to the ISU via an IOC control channel.
The mechanisms in the ISU which adjust the parameters for upstream
synchronization include implementing an amplitude change with a scalar
multiplication of the time domain waveform as it is being collected from
the digital processing algorithm, such as inverse FFT 190, by the digital
to analog converters 194 (FIG. 24). Similarly, a complex mixing signal
could be created and implemented as a complex multiply applied to the
input to the digital to analog converters 194.
Frequency accuracy of both the downstream sample clock and upstream sample
clock, in the ISU, is established by phase locking an oscillator to the
downstream synchronization and IOC information. Upstream transmission
frequency is adjusted, for example, at synthesizer block 195 as commanded
by the HDT 12.
Symbol timing corrections are implemented as a delay function. Symbol
timing alignment in the ISU upstream direction is therefore established as
a delay in the sample timing accomplished by either blanking a sample
interval (two of the same samples to go out simultaneously) or by putting
in an extra clock edge (one sample is clocked out and lost) via clock
delay 196 (FIG. 24). In this manner, a delay function can be controlled
without data storage overhead beyond that already required.
After the ISU is initialized and activated into the system, ready for
transmission, the ISU will maintain required upstream synchronization
system parameters using the carrier, amplitude, frequency recovery block
222. An unused but initialized and activated ISU will be commanded to
transmit on an IOC and the block 222 will estimate the parameters
therefrom as explained above.
In both the upstream transmitter architectures for the MISU 66 (FIG. 24)
and the HISU 68 (FIG. 25), frequency offset or correction to achieve
orthogonality of the carriers at HDT 12 can be determined on the ISU as
opposed to the frequency offset being determined at the HDT during
synchronization by carrier, amplitude and timing recovery block 222 (FIG.
26) and then frequency offset adjustment commands being transmitted to the
ISU for adjustment of carrier frequency via the synthesizer blocks 195 and
199 of the HISU 68 and MISU 66, respectively. Thus, frequency error would
no longer be detected by carrier, amplitude and timing recovery block 222
as described above. Rather, in such a direct ISU implementation, the ISU,
whether an HISU 68 or MISU 66, estimates a frequency error digitally from
the downstream signal and a correction is applied to the upstream data
being transmitted.
The HDT 12 derives all transmit and receive frequencies from the same
fundamental oscillator. Therefore, all mixing signals are frequency locked
in the HDT. Similarly, the ISU, whether an HISU 68 or MISU 66, derives all
transmit and receive frequencies from the same fundamental oscillator;
therefore, all the mixing signals on the ISU are also frequency locked.
There is, however, a frequency offset present in the ISU oscillators
relative to the HDT oscillators. The amount of frequency error (viewed
from the ISU) will be a fixed percentage of the mixing frequency. For
example, if the ISU oscillator is 10 PPM off in frequency from the HDT
oscillators, and the downstream ISU receiver mix frequency was 100 MHz and
the ISU upstream transmit mixing frequency were 10 MHz, the ISU would have
to correct for 1 kHz on the downstream receiver and create a signal with a
100 Hz offset on the upstream transmitter. As such, with the ISU direct
implementation, the frequency offset is estimated from the downstream
signal.
The estimation is performed with digital circuitry performing numeric
calculations, i.e. a processor. Samples of the synchronization channel or
IOC channel are collected in hardware during operation of the system. A
tracking loop drives a digital numeric oscillator which is digitally mixed
against the received signal. This process derives a signal internally that
is essentially locked to the HDT. The internal numerical mix accounts for
the frequency offset. During the process of locking to the downstream
signal in the ISU, the estimate of frequency error is derived and with the
downstream frequency being known, a fractional frequency error can be
computed. Based on the knowledge of the mixing frequency at the HDT that
will be used to down convert the upstream receive signal, an offset to the
ISU transmit frequency is computed. This frequency offset is digitally
applied to the ISU transmitted signal prior to converting the signal to
the analog domain, such as by converters 194 of FIG. 24. Therefore, the
frequency correction can be performed directly on the ISU.
Referring to FIGS. 31 and 32, the narrow band ingress filter and FFT 112 of
the MCC upstream receiver architecture, including a polyphase filter
structure, will be described in further detail. Generally, the polyphase
filter structure includes polyphase filters 122 and 124 and provides
protection against ingress. The 6 MHz band of upstream OFDM carriers from
the ISUs 100 is broken into subbands through the polyphase filters which
provide filtering for small groups of carriers or tones, and if an ingress
affects carriers within a group of carriers, only that group of carriers
is affected and the other groups of carriers are protected by such
filtering characteristics.
The ingress filter structure has two parallel banks 122, 124 of polyphase
filters. One bank has approximately 17 different non-overlapping bands
with channel spaces between the bands. A magnitude response of a single
polyphase filter bank is shown in FIG. 29. The second bank is offset from
the first bank by an amount so that the channels that are not filtered by
the first bank are filtered by the second bank. Therefore, as shown in the
closeup magnitude response of a single polyphase filter bank in FIG. 30,
one band of channels filtered may include those in frequency bins 38-68
with the center carriers corresponding to bins 45-61 being passed by the
filter. The overlapping filter provides for filtering carriers in the
spaces between the bands and the carriers not passed by the other filter
bank. For example, the overlapping filter may pass 28-44. The two channel
banks are offset by 16 frequency bins so that the combination of the two
filter banks receives every one of the 544 channels.
Referring to FIG. 31, the ingress filter structure receives the sampled
waveform x(k) from the analog to digital converters 212 and then complex
mixers 118 and 120 provide the stagger for application to the polyphase
filters 122, 124. The mixer 118 uses a constant value and the mixer 120
uses a value to achieve such offset The outputs of each mixer enters one
of the polyphase filters 122, 124. The output of each polyphase filter
bank comprises 18 bands, each of which contain 16 usable FFT bins or each
band supports sixteen carriers at the 8 kHz rate, or 8 DS0+s. One band is
not utilized.
Each band output of the polyphase filters 122, 124 has 36 samples per 8 kHz
frame including 4 guard samples and enters a Fast Fourier Transform (FFT)
block 126, 128. The first operation performed by the FFT blocks 126, 128
is to remove the four guard samples, thereby leaving 32 time domain
points. The output of the each FFT in the blocks is 32 frequency bins, 16
of which are used with the other bins providing filtering. The output of
the FFTs are staggered to provide overlap. As seen in FIG. 31, carriers
0-15 are output by FFT #1 of the top bank, carriers 16-31 are output by
FFT #1 of the bottom bank, carriers 32-48 are output by FFT #2 of the top
bank and so on.
The polyphase filters 122, 124 are each standard polyphase filter
construction as is known to one skilled in the art and each is shown by
the structure of FIG. 32. The input signal is sampled at a 5.184
mega-sample per second rate, or 648 samples per frame. The input is then
decimated by a factor of 18 (1 of 18 samples are kept) to give an
effective sample rate of 288 kHz. This signal is subjected to the finite
impulse response (FIR) filters, labeled H.sub.0,0 (Z) through H.sub.0,16
(Z), which include a number of taps, preferably 5 taps per filter. As one
skilled in the art will recognize the number of taps can vary and is not
intended to limit the scope of the invention. The outputs from the filters
enter an 18 point inverse FFT 130. The output of the transform is 36
samples for an 8 kHz frame including 4 guard samples and is provided to
FFT blocks 126 and 128 for processing as described above. The FFT tones
are preferably spaced at 9 kHz, and the information rate is 8 kilosymbols
per second with four guard samples per symbol allotted. The 17 bands from
each polyphase filter are applied to the FFT blocks 126, 128 for
processing and output of the 544 carriers as indicated above. One band,
the 18th band, as indicated above, is not used.
The equalizer 214 (FIG. 26) and 172 (FIG. 22), in both upstream and
downstream receiver architectures, is supplied to account for changes in
group delay across the cable plant. The equalizer tracks out phase and
gain or amplitude variations due to environmental changes and can
therefore adapt slowly while maintaining sufficiently accurate tracking.
The coefficients 360 of the equalizer 172, 214, for which the internal
equalizer operation is generally shown in FIG. 35, represent the inverse
of the channel frequency response to the resolution of the FFT 112, 170.
The downstream coefficients will be highly correlated since every channel
will progress through the same signal path as opposed to the upstream
coefficients which may be uncorrelated due to the variant channels that
individual DS0+s may encounter in the multi-point to point topology. While
the channel characteristics are diverse, the equalizer will operate the
same for either upstream or downstream receiver.
The downstream equalizer will track on only the IOC channels, thus reducing
the computational requirements at the ISUs and removing the requirement
for a preamble in the payload channels, as described further below, since
the IOC channels are always transmitted. The upstream, however, will
require equalization on a per DS0+ and IOC channel basis.
The algorithm used to update the equalizer coefficients contains several
local minima when operating on a 32 QAM constellation and suffers from a
four-fold phase ambiguity. Furthermore, each DS0+ in the upstream can
emanate from a separate ISU, and can therefore have an independent phase
shift. To mitigate this problem, each communication onset will be required
to post a fixed symbol preamble prior to data transmission. Note that the
IOC channels are excluded from this requirement since they are not
equalized and that the preamble cannot be scrambled. It is known that at
the time of transmission, the HDT 12 will still have accurate frequency
lock and symbol timing as established during initialization and activation
of the ISU and will maintain synchronization on the continuously available
downstream IOC channel.
The introduction of the preamble requires that the equalizer have knowledge
of its process state. Three states are introduced which include: search,
acquisition, and tracking mode. Search mode is based on the amount of
power present on a channel. Transmitter algorithms will place a zero value
in unused FFT bins, resulting in no power being transmitted on that
particular frequency. At the receiver, the equalizer will determine that
it is in search mode based on the absence of power in the FFT bin.
When transmission begins for an initialized and activated ISU, the
equalizer detects the presence of signal and enters the acquisition mode.
The length of the preamble may be about 15 symbols. The equalizer will
vary the equalization process based on the preamble. The initial phase and
amplitude correction will be large but subsequent updates of the
coefficients will be less significant. In order to differentiate the
training pattern from any other data sequence, when the HDT informs an ISU
to connect a new payload channel, the ISU transmits 16 consecutive symbols
having 1=0 and Q=0, which is not a valid data symbol in the constellations
of FIGS. 14 or 17. The ISU then transmits 8 valid data symbols, allowing
the equalizer for that channel to set its coefficient properly to adjust
for amplitude and phase of the incoming signal.
After acquisition, the equalizer will enter a tracking mode with the update
rate being reduced to a minimal level. The tracking mode will continue
until a loss of power is detected on the channel for a period of time. The
channel is then in the unused but initialized and activated state. The
equalizer will not train or track when the receiver is being tuned and the
coefficients will not be updated. The coefficients may be accessed and
used such as by signal to noise detector 305 (FIG. 26) for channel
monitoring as discussed further below.
For the equalization process, the I/Q components are loaded into a buffer
at the output of the FFT, such as FFT 112, 180. As will be apparent to one
skilled in the art, the following description of the equalizer structure
is with regard to the upstream receiver equalizer 214 but is equally
applicable to the downstream receiver equalizer 172. The equalizer 214
extracts time domain samples from the buffer and processes one complex
sample at a time. The processed information is then output therefrom. FIG.
35 shows the basic structure of the equalizer algorithm less the state
control algorithm which should be apparent to one skilled in the art. The
primary equalization path performs a complex multiply at multiplier 370
with the value from the selected FFT bin. The output is then quantized at
symbol quantize block 366 to the nearest symbol value from a storage
table. The quantized value (hard decision) is passed out to be decoded
into bits by symbols to bits converter 216. The remainder of the circuitry
is used to update the equalizer coefficients. An error is calculated
between the quantized symbol value and the equalized sample at summer 364.
This complex error is multiplied with the received sample at multiplier
363 and the result is scaled by the adaptation coefficient by multiplier
362 to form an update value. The update value is summed at summer 368 with
the original coefficient to result in a new coefficient value.
Operation of First Embodiment
In the preferred embodiment, the 6 MHz frequency band for each MCC modem 82
of HDT 12 is allocated as shown in FIG. 13. Although the MCC modem 82
transmits and receives the entire 6 MHz band, the ISU modems 100 (FIG. 8)
are optimized for the specific application for which they are designed and
may terminate/generate fewer than the total number of carriers or tones
allocated in the 6 MHz band. The upstream and downstream band allocations
are preferably symmetric. The upstream 6 MHz bands from the MCC modems 82
lie in the 5-40 MHz spectrum and the downstream 6 MHz bands lie in the
725-760 MHz spectrum. One skilled in the art will recognize that if
different transmission media are utilized for upstream and downstream
transmission, the frequencies for transmission may be the same or overlap
but still be non-interfering.
There are three regions in each 6 MHz frequency band to support specific
operations, such as transport of telephony payload data, transport of ISU
system operations and control data (IOC control data), and upstream and
downstream synchronization. Each carrier or tone in the OFDM frequency
band consists of a sinusoid which is modulated in amplitude and phase to
form a complex constellation point as previously described. The
fundamental symbol rate of the OFDM waveform is 8 kHz, and there are a
total of 552 tones in the 6 MHz band. The following Table 2 summarizes the
preferable modulation type and bandwidth allocation for the various tone
classifications.
TABLE 2
Number of
Band Tones or
Allocation Carriers Modulation Capacity Bandwidth
Synch 24 tones(2 synch BPSK n/a 216 kHz
Band tones at each end
and 10 guard
tones at each end)
Payload 480 (240 DSO + 32 QAM 19.2 MBPS 4.32 MHz
Data channels)
IOC 48 (2 every 20 BPSK 384 kbps 432 kHz
data channels or
24 IOC channels)
Intra-band Remainder on n/a n/a 1.032 MHz
guard each end (516 kHz at
each end)
Composite 552 n/a n/a 6.0 MHz
Signal
Guard bands are provided at each end of the spectrum to allow for
selectivity filtering after transmission and prior to reception. A total
of 240 telephony data channels are included throughout the band, which
accommodates a net data rate of 19.2 Mbps. This capacity was designed to
account for additive ingress, thereby retaining enough support to achieve
concentration of users to the central office. The IOC channels are
interspersed throughout the band to provide redundancy and communication
support to narrowband receivers located in the HISUs. The IOC data rate is
16 kbps (two BPSK tones at the symbol rate of 8 kHz frames per second).
Effectively, an IOC is provided for every 10 payload data channels. An
ISU, such as an HISU, that can only see a single IOC channel would be
forced to retune if the IOC channel is corrupted. However, an ISU which
can see multiple IOC channels can select an alternate IOC channel in the
event that the primary choice is corrupt, such as for an MISU.
The synchronization channels are duplicated at the ends of the band for
redundancy, and are offset from the main body of usable carriers to
guarantee that the synchronization channels do not interfere with the
other used channels. The synchronization channels were previously
described and will be further described below. The synchronization
channels are operated at a lower power level than the telephony payload
channels to also reduce the effect of any interference to such channels.
This power reduction also allows for a smaller guard band to be used
between the synchronization channels and the payload telephony channels.
One synchronization or redundant synchronization channels may also be
implemented within the telephony channels as opposed to being offset
therefrom. In order to keep them from interfering with the telephony
channels, the synchronization channels may be implemented using a lower
symbol rate. For example, if the telephony channels are implemented at an
8 kHz symbol rate, the synchronization channels could be implemented at a
2 kHz symbol rate and also may be at a lower power level.
The ISUs 100 are designed to receive a subband, as shown in FIG. 16, of the
total aggregate 6 MHz spectrum. As an example, the HISU 68 will preferably
detect only 22 of the available 552 channels. This implementation is
primarily a cost/power savings technique. By reducing the number of
channels being received, the sample rate and associated processing
requirements are dramatically reduced and can be achievable with common
conversion parts on the market today.
A given HISU 68 is limited to receiving a maximum of 10 DS0s out of the
payload data channels in the HISU receiver's frequency view. The remaining
channels will be used as a guard interval. Furthermore, in order to reduce
the power/cost requirements, synthesizing frequency steps will be limited
to 198 kHz. An IOC channel is provided for as shown in FIG. 16 so that
every HISU 68 will always see an IOC channel for control of the HISU 68
via HDT 12.
The MISU 66 is designed to receive 13 subbands, as shown in FIG. 16, or 130
of the 240 available DS0s. Again, the tuning steps will be limited to 198
kHz to realize an efficient synthesizer implementation. These are
preferred values for the HISU 68 and MISU 66, and it will be noted by one
skilled in the art that many of the values specified herein can be varied
without changing the scope or spirit of the invention as defined by the
accompanying claims.
As known to one skilled in the art, there may be need to support operation
over channels in a bandwidth of less than 6 MHz. With appropriate software
and hardware modifications of the system, such reconfiguration is possible
as would be apparent to one skilled in the art. For example, for a 2 MHz
system, in the downstream, the HDT 12 would generate the channels over a
subset of the total band. The HISUs are inherently narrowband and would be
able to tune into the 2 MHz band. The MISUs supporting 130 channels, would
receive signals beyond the 2 MHz band. They would require reduction in
filter selectivity by way of a hardware modification. An eighty (80)
channel MISU would be able to operate with the constraints of the 2 MHz.
system. In the upstream, the HISUs would generate signals within the 2 MHz
band and the MISUs transmit section would restrict the information
generated to the narrower band. At the HDT, the ingress filtering would
provide sufficient selectivity against out of band signal energy. The
narrowband system would require synchronization bands at the edges of the
2 MHz band.
As previously described, acquisition of signal parameters for initializing
the system for detection of the downstream information is performed using
the downstream synchronization channels. The ISUs use the carrier,
amplitude, timing recovery block 166 to establish the downstream
synchronization of frequency, amplitude and timing for such detection of
downstream information. The downstream signal constitutes a point to
multi-point topology and the OFDM waveform arrives at the ISUs via a
single path in an inherently synchronous manner. In the upstream
direction, each ISU 100 must be initialized and activated through a
process of upstream synchronization before an HDT 12 can enable the ISU
100 for transmission. The process of upstream synchronization for the ISUs
is utilized so that the waveform from distinct ISUs combine to a unified
waveform at the HDT. The upstream synchronization process, portions of
which were previously described, involves various steps. They include: ISU
transmission level adjustment, upstream multicarrier symbol alignment,
carrier frequency adjustment, and round trip path delay adjustment. Such
synchronization is performed after acquisition of a 6 MHz band of
operation.
Generally, with respect to level adjustment, the HDT 12 calibrates the
measured signal strength of the upstream transmission received from an ISU
100 and adjusts the ISU 100 transmit level so that all ISUs are within
acceptable threshold. Level adjustment is performed prior to symbol
alignment and path delay adjustment to maximize the accuracy of these
measurements.
Generally, symbol alignment is a necessary requirement for the multicarrier
modulation approach implemented by the MCC modems 82 and the ISU modems
101. In the downstream direction of transmission, all information received
at the ISU 100 is generated by a single CXMU 56, so the symbols modulated
on each multicarrier are automatically phase aligned. However, upstream
symbol alignment at the MCC modem 82 receiver architecture varies due to
the multi-point to point nature of the HFC distribution network 11 and the
unequal delay paths of the ISUs 100. In order to have maximum receiver
efficiency, all upstream symbols must be aligned within a narrow phase
margin. This is done by providing an adjustable delay path parameter in
each ISU 100 such that the symbol periods of all channels received
upstream from the different ISUs are aligned at the point they reach the
HDT 12.
Generally, round trip path delay adjustment is performed such that the
round trip delay from the HDT network interface 62 to all ISUs 100 and
back to the network interface 62 from all the ISUs 100 in a system must be
equal. This is required so that signaling multiframe integrity is
preserved throughout the system. All round trip processing for the
telephony transport section has a predictable delay with the exception of
the physical delay associated with signal propagation across the HFC
distribution network 11 itself. ISUs 100 located at close physical
distance from the HDT 12 will have less round trip delay than ISUs located
at the maximum distance from the HDT 12. Path delay adjustment is
implemented to force the transport system of all ISUs to have equal round
trip propagation delay. This also maintains DS1 multiframe alignment for
DS1 channels transported through the system, maintaining in-band channel
signaling or robbed-bit signaling with the same alignment for voice
services associated with the same DS1.
Generally, carrier frequency adjustment must be performed such that the
spacing between carrier frequencies is such as to maintain orthogonality
of the carriers. If the multicarriers are not received at the MCC modem 82
in orthogonal alignment, interference between the multicarriers may occur.
Such carrier frequency adjustment can be performed in a manner like that
of symbol timing or amplitude adjustment or may be implemented on the ISU
as described previously above.
In the initialization process, when the ISU has just been powered up, the
ISU 100 has no knowledge of which downstream 6 MHz frequency band it
should be receiving in. This results in the need for the acquisition of 6
MHz band in the initialization process. Until an ISU 100 has successfully
acquired a 6 MHz band for operation, it implements a "scanning" approach
to locate its downstream frequency band. Generally, a local processor of
the CXSU controller 102 of ISU 100 starts with a default 6 MHz receive
frequency band somewhere in the range from 625 to 850 MHz. The ISU 100
waits for a period of time, for example 100 milliseconds, in each 6 MHz
band to look for a valid 6 MHz acquisition command which matches a unique
identification number for the ISU 100 after obtaining a valid
synchronization signal; which unique identifier may take the form of or be
based on a serial number of the ISU equipment. If a valid 6 MHz
acquisition command or valid synchronization command is not found in that
6 MHz band, the CXSU controller 102 looks at the next 6 MHz band and the
process is repeated. In this manner, as explained further below, the HDT
12 can tell the ISU 100 which 6 MHz band it should use for frequency
reception and later which band for frequency transmission upstream.
The process of initialization and activation of ISUs, as generally
described above, and tracking or follow-up synchronization is further
described below. This description is written using an MISU 66 in
conjunction with a CXSU controller 102 but is equally applicable to any
ISU 100 implemented with an equivalent controller logic. The coax master
card logic (CXMC) 80 is instructed by the shelf controller unit (SCNU) 58
to initialize and activate a particular ISU 100. The SCNU uses an ISU
designation number to address the ISU 100. The CXMC 80 correlates the ISU
designation number with an equipment serial number, or unique identifier,
for the equipment. No two ISU equipments shipped from the factory possess
the same unique identifier. If the ISU 100 has never before been
initialized and activated in the current system database, the CXMC 80
chooses a personal identification number (PIN) code for the ISU 100 being
initialized and activated. This PIN code is then stored in the CXMC 80 and
effectively represents the "address" for all communications with that ISU
100 which will follow. The CXMC 80 maintains a lookup table between each
ISU designation number, the unique identifier for the ISU equipment, and
the PIN code. Each ISU 100 associated with the CXMU 56 has a unique PIN
address code assignment. One PIN address code will be reserved for a
broadcast feature to all ISUs, which allows for the HDT to send messages
to all initialized and activated ISUs 100.
The CXMC 80 sends an initialization and activation enabling message to the
MCC modem 82 which notifies the MCC modem 82 that the process is beginning
and the associated detection functionality in the MCC modem 82 should be
enabled. Such functionality is performed at least in part by carrier,
amplitude, timing recovery block 222 as shown in the MCC upstream receiver
architecture of FIG. 26 and as previously discussed.
The CXMC 80 transmits an identification message by the MCC modem 82 over
all IOC channels of the 6 MHz band in which it transmits. The message
includes a PIN address code to be assigned to the ISU being initialized
and activated, a command indicating that ISU initialization and activation
should be enabled at the ISU 100, the unique identifier for the ISU
equipment, such as the equipment serial number, and cyclical redundancy
checksum (CRC). The messages are sent periodically for a certain period of
time, T.sub.SCAN, which is shown as 6.16 seconds in FIG. 20 and which is
also shown in FIG. 19. This period of time is the maximum time which an
ISU can scan all downstream 6 MHz bands, synchronize, and listen for a
valid identification message. The periodic rate, for example 50 msec,
affects how quickly the ISU learns its identity. The CXMC 80 will never
attempt to synchronize more than one ISU at a time, but will attempt to
identify several ISUs during burst identification as described further
below. A software timeout is implemented if an ISU does not respond after
some maximum time limit is exceeded. This timeout must be beyond the
maximum time limit required for an ISU to obtain synchronization
functions.
During periodic transmission by CXMC 80, the ISU implements the scanning
approach to locate its downstream frequency band. The local processor of
the CXSU starts with a default 6 MHz receive frequency band somewhere in
the range from 625 to 850 MHz. The ISU 100 selects the primary
synchronization channel of the 6 MHz band and then tests for loss of
synchronization after a period of time. If loss of synchronization is
still present, the secondary synchronization channel is selected and
tested for loss of synchronization after a period of time. If loss of
synchronization is still present, then the ISU restarts selection of the
synchronization channels on the next 6 MHz band which may be 1 MHz away
but still 6 MHz in width. When loss of synchronization is not present on a
synchronization channel then the ISU selects the first subband including
the IOC and listens for a correct identification message. If a correct
identification message is found which matches its unique identifier then
the PIN address code is latched into an appropriate register. If a correct
identification message is not found in the first subband on that IOC then
a middle subband and IOC is selected, such as the 11th subband, and the
ISU again listens for the correct identification message. If the message
again is not correctly detected, then the ISU restarts on another 6 MHz
band. The ISU listens for the correct identification message in a subband
for a period of time equal to at least two times the CXMU transmission
time, for example 100 msec when transmission time is 50 msec as described
above. The identification command is a unique command in the ISU 100, as
the ISU 100 will not require a PIN address code match to respond to such
commands, but only a valid unique identifier and CRC match. If an
un-initialized and un-activated ISU 100 receives an identification command
from the CXMC 80 via the MCC modem 82 on an IOC channel, data which
matches the unique identifier and a valid CRC, the CXSU 102 of the ISU 100
will store the PIN address code transmitted with the command and unique
identifier. From this point on, the ISU 100 will only respond to commands
which address it by its correct PIN address code, or a broadcast address
code; unless, of course, the ISU is re-activated again and given a new PIN
address code.
After the ISU 100 has received a match to its unique identifier, the ISU
will receive the upstream frequency band command with a valid PIN address
code that tells the ISU 100 which 6 MHz band to use for upstream
transmission and the carrier or tone designations for the upstream IOC
channel to be used by the ISU 100. The CXSU controller 102 interprets the
command and correctly activates the ISU modem 101 of the ISU 100 for the
correct upstream frequency band to respond in. Once the ISU modem 101 has
acquired the correct 6 MHz band, the CXSU controller 102 sends a message
command to the ISU modem 101 to enable upstream transmission. Distributed
loops utilizing the carrier, amplitude, and timing recovery block 222 of
the MCC modem upstream receiver architecture of the HDT 12 is used to lock
the various ISU parameters for upstream transmission, including amplitude,
carrier frequency, symbol alignment, and path delay.
The HDT is then given information on the new ISU and provides downstream
commands for the various parameters to the subscriber ISU unit The ISU
begins transmission in the upstream and the HDT 12 locks to the upstream
signal. The HDT 12 derives an error indicator with regard to the parameter
being adjusted and commands the subscriber ISU to adjust such parameter.
The adjustment of error is repeated in the process until the parameter for
ISU transmission is locked to the HDT 12.
More specifically, after the ISU 100 has acquired the 6 MHz band for
operation, the CXSU 102 sends a message command to the ISU modem 101 and
the ISU modem 101 transmits a synchronization pattern on a synchronization
channel in the primary synchronization band of the spectral allocation as
shown in FIGS. 13-18. The upstream synchronization channels which are
offset from the payload data channels as allocated in FIGS. 13-18 include
both a primary and a redundant synchronization channel such that upstream
synchronization can still be accomplished if one of the synchronization
channels is corrupted. The HDT monitors one channel for every ISU.
The MCC modem 82 detects a valid signal and performs an amplitude level
measurement on a received signal from the ISU. The synchronization pattern
indicates to the CXMC 80 that the ISU 100 has received the activation and
initialization and frequency band commands and is ready to proceed with
upstream synchronization. The amplitude level is compared to a desired
reference level. The CXMC 80 determines whether or not the transmit level
of the ISU 100 should be adjusted and the amount of such adjustment If
level adjustment is required, the CXMC 80 transmits a message on the
downstream IOC channel instructing the CXSU 102 of the ISU 100 to adjust
the power level of the transmitter of the ISU modem 101. The CXMC 80
continues to check the receive power level from the ISU 100 and provides
adjustment commands to the ISU 100 until the level transmitted by the ISU
100 is acceptable. The amplitude is adjusted at the ISU as previously
discussed. If amplitude equilibrium is not reached within a certain number
of iterations of amplitude adjustment or if a signal presence is never
detected utilizing the primary synchronization channel then the same
process is used on the redundant synchronization channel. If amplitude
equilibrium is not reached within a certain number of iterations of
amplitude adjustment or if a signal presence is never detected utilizing
the primary or redundant synchronization channels then the ISU is reset.
Once transmission level adjustment of the ISU 100 is completed and has been
stabilized, the CXMC 80 and MCC modem 82 perform carrier frequency
locking. The MCC modem 82 detects the carrier frequency as transmitted by
the ISU 100 and performs a correlation on the received transmission from
the ISU to calculate a carrier frequency error correction necessary to
orthogonally align the multicarriers of all the upstream transmissions
from the ISUs. The MCC modem 82 returns a message to the CXMC 80
indicating the amount of carrier frequency error adjustment required to
perform frequency alignment for the ISU. The CXMC 80 transmits a message
on a downstream IOC channel via the MCC modem 82 instructing the CXSU 102
to adjust the transmit frequency of the ISU modem 101 and the process is
repeated until the frequency has been established to within a certain
tolerance for the OFDM channel spacing. Such adjustment would be made via
at least the synthesizer block 195 (FIGS. 24 and 25). If frequency locking
and adjustment is accomplished on the ISU as previously described, then
this frequency adjustment method is not utilized.
To establish orthogonality, the CXMC 80 and MCC modem 82 perform symbol
alignment. The MCC modem 82 detects the synchronization channel modulated
at a 8 kHz frame rate transmitted by the ISU modem 101 and performs a
hardware correlation on the receive signal to calculate the delay
correction necessary to symbol align the upstream ISU transmission from
all the distinct ISUs 100. The MCC modem 82 returns a message to the CXMC
80 indicating the amount of delay adjustment required to symbol align the
ISU 100 such that all the symbols are received at the HDT 12
simultaneously. The CXMC 80 transmits a message in a downstream IOC
channel by the MCC modem 82 instructing the CXSU controller 102 to adjust
the delay of the ISU modem 101 transmission and the process repeats until
ISU symbol alignment is achieved. Such symbol alignment would be adjusted
via at least the clock delay 196 (FIGS. 24 and 25). Numerous iterations
may be necessary to reach symbol alignment equilibrium and if it is not
reached within a predetermined number of iterations, then the ISU may
again be reset.
Simultaneously with symbol alignment, the CXMC 80 transmits a message to
the MCC modem 82 to perform path delay adjustment. The CXMC 80 sends a
message on a downstream IOC channel via the MCC modem 82 instructing the
CXSU controller 102 to enable the ISU modem 101 to transmit a another
signal on a synchronization channel which indicates the multiframe (2 kHz)
alignment of the ISU 100. The MCC modem 82 detects this multiframe
alignment pattern and performs a hardware correlation on the pattern. From
this correlation, the modem 82 calculates the additional symbol periods
required to meet the round trip path delay of the communication system.
The MCC modem 82 then returns a message to the CXMC 80 indicating the
additional amount of delay which must be added to meet the overall path
delay requirements and the CXMC then transmits a message on a downstream
IOC channel via the MCC modem 82 instructing the CXSU controller 102 to
relay a message to the ISU modem 101 containing the path delay adjustment
value. Numerous iterations may be necessary to reach path delay
equilibrium and if it is not reached within a predetermined number of
iterations, then the ISU may again be reset. Such adjustment is made in
the ISU transmitter as can be seen in the display delay buffer "n" samples
192 of the upstream transmitter architectures of FIGS. 24 and 25. Path
delay and symbol alignment may be performed at the same time, separately
or together using the same or different signals sent on the
synchronization channel.
Until the ISU is initialized and activated, the ISU 100 has no capability
of transmitting telephony data information on any of the 480 tones or
carriers. After such initialization and activation has been completed, the
ISUs are within tolerance required for transmission within the OFDM
waveform and the ISU is informed that transmission is possible and
upstream synchronization is complete.
After an ISU 100 is initialized and activated for the system, follow-up
synchronization or tracking may be performed periodically to keep the ISUs
calibrated within the required tolerance of the OFDM transport
requirements. The follow-up process is implemented to account for drift of
component values with temperature. If an ISU 100 is inactive for extreme
periods of time, the ISU can be tuned to the synchronization channels and
requested to update upstream synchronization parameters in accordance with
the upstream synchronization process described above. Alternatively, if an
ISU has been used recently, the follow-up synchronization or tracking can
proceed over an IOC channel. Under this scenario, as generally shown in
FIG. 28, the ISU 100 is requested to provide a signal over an IOC channel
by the HDT 12. The HDT 12 then acquires and verifies that the signal is
within the tolerance required for a channel within the OFDM waveform. If
not, then the ISU is requested to adjust such errored parameters. In
addition, during long periods of use, ISUs can also be requested by the
HDT 12 to send a signal on an IOC channel or a synchronization channel for
the purpose of updating the upstream synchronization parameters.
In the downstream direction, the IOC channels transport control information
to the ISUs 100. The modulation format is preferably differentially
encoded BPSK, although the differential aspect of the downstream
modulation is not required. In the upstream direction, the IOC channels
transport control information to the HDT 12. The IOC channels are
differentially BPSK modulated to mitigate the transient time associated
with the equalizer when sending data in the upstream direction. Control
data is slotted on a byte boundary (500 .mu.s frame). Data from any ISU
can be transmitted on an IOC channel asynchronously; therefore, there is
the potential for collisions to occur.
As there is potential for collisions, detection of collisions on the
upstream IOC channels is accomplished at a data protocol level. The
protocol for handling such collisions may, for example, include
exponential backoff by the ISUs. As such, when the HDT 12 detects an error
in transmission, a retransmission command is broadcast to all the ISUs
such that the ISUs retransmit the upstream signal on the IOC channel after
waiting a particular time; the wait time period being based on an
exponential function.
One skilled in the art will recognize that upstream synchronization can be
implemented allowing for multi-point to point transmission using only the
symbol timing loop for adjustment of symbol timing by the ISUs as
commanded by the HDT. The frequency loop for upstream synchronization can
be eliminated with use of high quality local free running oscillators in
the ISUs that are not locked to the HDT. In addition, the local
oscillators at the ISUs could be locked to an outside reference. The
amplitude loop is not essential to achieve symbol alignment at the HDT.
In the process described above with respect to initialization and
activation, including upstream synchronization, if for some reason
communication is lost between a large number of ISUs 100 and the HDT 12,
after a period of time these ISUs 100 will require initialization and
activation once again. Such a case may arise when a fiber is cut and users
of multiple ISUs 100 are left without service. As initialization and
activation is described above, only one ISU 100 would be initialized and
activated at one time. The time frame for initialization and activation of
multiple ISUs 100 in this manner is shown in FIG. 19.
In FIG. 19, each ISU 100 is initialized, as previously described, by
identification of the ISU and acquisition by the ISU of the 6 MHz band for
downstream transmission during a scanning period T.sub.SCAN, which is the
time period needed for the ISU to scan all of the downstream bands
listening for a matching identification message. In one embodiment,
T.sub.SCAN is equal to 6.16 seconds. Of course this time period is going
to be dependant on the number of bands scanned, the time period necessary
for synchronizing on the downstream synchronization channels, and the time
required to acquire an IOC channel within the band.
Further as shown in FIG. 19, after each ISU has acquired a downstream and
upstream 6 MHz band, upstream synchronization is then performed during a
time period T.sub.EQUAL. T.sub.EQUAL may be defined as the period in which
an ISU should have received all messages from the CXMC 80 finishing the
upstream synchronization process as described above, with a reasonable
amount of iterations to accomplish such synchronization. At the very
least, this time period is the time period necessary to accomplish symbol
timing such that symbols received from various ISUs 100 at the HDT 12 are
orthogonal. This time period would be increased if amplitude, frequency
and path delay synchronization is also performed as described above.
Therefore, the time period necessary to serially initialize and activate
twelve ISUs, T.sub.SERIAL, as shown in FIG. 19 would be equal to
12T.sub.SCAN +12T.sub.EQUAL.
With a burst identification process as shall be described with respect to
FIG. 20, the time period for initializing and activating twelve ISUs 100
can be substantially reduced. This results in more ISUs 100 being
activated more quickly and more users once again serviced in a shorter
period of time. In the process of burst identification as shown by the
timing diagram of FIG. 20, the identification and acquisition of multiple
ISUs 100 is performed in parallel instead of being performed serially as
described above.
The periodicity of the identification messages sent by the CXMC 80 during
initialization and acquisition, when performed during normal operating
conditions when ISUs 100 are serially initialized, is designed to present
a light load of traffic on the IOC channel but yet to allow a reasonable
identification duration. This periodicity duration is, for example, 50
milliseconds. For the system to be able to handle both situations, serial
identification and burst identification, this periodicity is kept the
same. However, in burst identification, the IOC channel traffic load is
not important because the service of all the ISUs 100 receiving
communication via one of the CXMCs 80 utilizing the IOC channels has been
terminated such as by the cut fiber. Therefore, during burst
identification the IOC channels can be loaded more heavily and
identification messages for multiple ISUs 100 utilizing such IOC channels
are transmitted on the IOC channels at the same periodicity as during
serial identification, but the phase for the identification messages is
different for each ISU.
Due to the period, and utilization of the IOC channels for the
identification messages during burst identification, there is a limit on
the number of identification messages which may be transmitted during one
T.sub.SCAN period. If the periodicity is 50 milliseconds and the use of
the IOC channel for a single identification message is 4 milliseconds,
only twelve ISUs 100 may be identified during one T.sub.SCAN period during
burst identification. As described below with further reference to FIG.
20, if the number of ISUs 100 desired to be burst identified is greater
than twelve, then multiple groups of burst identifications are serially
performed.
One skilled in the art will recognize that the specified numbers for time
periods are for illustration only and that the present invention is not
limited to such specified time periods. For example, the periodicity may
be 100 milliseconds and the number of ISUs identified during burst
identification may be 24. Many different time periods may be specified
with corresponding changes made to the other times periods specified.
Further, burst identification may be accomplished having a periodicity
different than that for serial identification.
As shown by the timing diagram of FIG. 19, a single burst initialization
and activation of twelve inactive ISUs 100 can be accomplished in the time
period T.sub.BURST which is equal to T.sub.SCAN +12T.sub.EQUAL. This is an
11T.sub.SCAN difference from the process carried out serially. During the
T.sub.SCAN period, identification messages for all twelve ISUs 100 being
initialized are transmitted on the IOC channels for a CXMC 80. The twelve
identification messages are each sent once during each 50 millisecond
period. The phase of each message is however different. For example,
identification message for ISU0 may be sent at time 0 and then again at 50
milliseconds, whereas the identification message for ISU1 may be sent at
time 4 milliseconds and then again at 54 milliseconds and so forth.
After the ISUs 100 being initialized have been identified and acquisition
of the downstream 6 MHz band has occurred during T.sub.SCAN, then upstream
synchronization is performed in a serial manner with respect to each ISU
identified during T.sub.SCAN. The upstream synchronization for the ISUs is
performed during the time period equal to 12T.sub.EQUAL. The CXMC 80 would
start the upstream synchronization process in the same manner as described
above for each ISU identified in a serial manner. The CXMC 80 sends to the
ISU the upstream transmission band in which the ISU being synchronized is
to transmit within and enables the upstream synchronization process to
begin. The upstream synchronization process for an ISU has been described
in detail above. If an upstream transmission band is not received and
upstream synchronization is not enabled for an ISU during the
12T.sub.EQUAL time period, then the ISU is reset at the end of the
12T.sub.EQUAL period by a period of time equal to T.sub.SCAN
+12T.sub.EQUAL to possibly perform upstream synchronization in the next
12T.sub.EQUAL period. Once a burst identification period, T.sub.BURST, is
completed, the process may be started over again in a second T.sub.BURST
period as shown in FIG. 20 if additional ISUs 100 are to be initialized
and activated.
FIGS. 47 and 48 describe a control loop distributed through the system for
acquiring and tracking an ISU, according to another embodiment of the
invention. Loop 3900 shows steps executed by the ISU 66 or 68 at the left,
and those executed by HDT 12 at the right. Messages between these two
units are shown as horizontal dashed lines; the IOC channels carry these
messages.
Steps 3910 prepare the ISU to communicate with the HDT. Step 3911 reads a
prestored internal table 3912 indicating frequencies of the valid RF
downstream bands, along with other information. Next, step 3913 tunes the
ISU's narrow-band receive modem to the center of the first 6 MHz band in
table 3912. Step 3913a then fine-tunes it to one of the two sync channels
in that band; assume that this "primary" channel is the one at the lower
end of the downstream part of the band shown in FIG. 13. Step 3914
acquires the amplitude and frequency of this sync tone. Briefly, equalizer
172, FIG. 22 or 23, is adjusted to bring the out put of the FFT to about
12 dB below its upper limit. Recovery block 166 measures the time for ten
frames of the sync tone, and compares it to the 1.25 msec correct time
interval; the frequency of synthesizer 157 is then adjusted accordingly.
Rough timing is satisfactory at this point, because the control messages
used below are simple low-frequency binary-keyed signals. If sync cannot
be acquired, exit 3915 causes block 3913a to retune to a secondary sync
tone of the same 6 MHz band, the upper downstream tone of FIG. 13. If the
ISU also fails to synchronize to the secondary tone, exit 3916 causes
block 3913 to tune to the center of the next 6 MHz band in table 3912 and
attempt synchronization again. If all bands have been tried, block 3913
continues to cycle through the bands again.
When step 3914 has acquired a lock, steps 3920 listen for messages from the
HDT. Step 3921 reads an internal predetermined manufacturer's serial
number which uniquely identifies that ISU, and compresses it into a
shorter, more convenient format. Step 3922 fine-tunes the ISU modem to a
designated primary subband, such as subband 0 in FIG. 16.
Concurrently, steps 3930 begin a search for the new ISU. Step 3931 receives
an operator "ranging" command to connect an ISU having a particular serial
number. Step 3932 then continuously broadcasts a "PIN message" 3933 on all
IOC channels; this message contains the compressed form of the ISU serial
number and a shorter personal identification number (PIN) by which that
ISU will be known. Step 3923 in the ISU continuously receives all PIN
messages, and attempts to match the transmitted compressed serial number
with the number from table 3922a. If it fails to do so after a period of
time, exit 3924 caused step 3922 to retune to another designated subband,
such as subband 23 in FIG. 13, and try again. If no appropriate PIN
message is received on the secondary subband, exit 3925 returns to step
3913 to switch to another 6 MHz band. When step 3923 receives the correct
PIN message, step 3926 latches the PIN into the ISU to serve henceforth as
its address within the system. In some implementations, the full serial
number or other unique identifier of the physical ISU could serve directly
as an address. However, this number occupies many bytes; it would be
wasteful to transmit it every time a message is addressed to the ISU, or
even to use it for ranging. Its compressed form, two bytes long, serves as
a hash code which is practical to transmit in the continuous messages
3933. The PIN is only one byte long, since addresses need be unique only
within each 6 MHz band, and are practical to use for identifying the ISU
whenever the HDT needs to communicate with it. Message 3927 informs the
HDT that the ISU will respond to its PIN.
Steps 3940 set up the upstream communications from the ISU to the HDT.
After 3941 receives PIN confirmation 3927, step 3942 sends a designation
of the upstream frequency band to the ISU as an IOC message 3943. This
frequency may have been specified by the operator in step 3931, or may
have been generated by the HDT itself. Step 3944 tunes the ISU modem to
this 6 Mhz band, and returns a confirmation 3945. Step 3946 then
fine-tunes to the primary upstream sync channel of that band, such as the
lower one in FIG. 13. Step 3947 enables an HDT receiver on the designated
band.
Steps 3950 adjust the transmitted power of the ISU in the upstream
direction; in a multipoint-to-point-system, the power levels of all ISUs
must track each other in order to ensure orthogonality of the entire
signal received by the modem of FIG. 26. Step 3951 transmits a ranging
tone at an initial power level from the ISU on this sync channel, which is
sometimes called a ranging channel. At the HDT, step 3952 measures the
received power level, and block 3953 sends an IOC message 3954, causing
step 3955 at the ISU to adjust the power of its transmitter 200, FIG. 24,
if necessary. If this cannot be done, message 3956 causes step 3946 to
retune to a secondary ranging channel, such as the higher upstream sync
tone of FIG. 13, and causes the step 3947 to enable the secondary channel
at the HDT. If this loop also fails, exit 3957 reports a hard failure to
the system logic.
Blocks 3960 align the symbol or frame timing between the ISU and the HDT.
Step 3961 measures the phase of the ISU's ranging tone with respect to the
sync tone that the HDT is sending to all ISUs at all times. FIG. 11 shows
this signal, labeled "8 kHz frame clk" in FIG. 24. Step 3962 sends
messages 3963 as necessary to cause the ISU modem to adjust the timing of
its ranging tone, in step 3964. When this has completed, block 3965
detects whether or not groups of four frames are aligned correctly as
between HDT and ISU; this grouping delineates boundaries of IOC messages,
which are four frames long. The sync tones continually repeat a
differential BPSK pattern of 1010 0101 0101 0101 over a period of 16
frames; that is, each bit occupies 125 microseconds, the duration of one
frame. Thus, the space between the fourth and fifth bit, and between the
sixteenth and first, can mark the multiframe boundaries. If alignment is
incorrect, step 3966 sends message 3967, causing the ISU to bump the phase
of its "2 kHz superframe clk", FIG. 24. If step 3961 or step 3965 cannot
reach the correct phase after a certain number of steps, fail exits 3969
report a hard failure.
Steps 3980 complete the induction of the new ISU into the system. Step 3981
turns off the ranging tone at the ISU, tells the HDT at 3982 that it is
locked in, and returns to the subband at which it was operating in step
3944. Step 3983 requests preliminary configuration or capability data from
the ISU, in message 3984, whereupon step 3985 reads an internal table 3986
containing parameters indicating capabilities such as whether the ISU can
tune only odd or even channel numbers, and other physical limitations of
that particular modem. When message 3987 has communicated these
parameters, step 3988 selects a particular subband of 10 or 130 payload
channels (for an HISU or an MISU respectively). Message 3989 causes step
3990 to tune the ISU to the proper subband. Meanwhile, the HDT is
acquiring an IOC data-link (IDL) channel, as described hereinbelow, at
step 3991. Step 3992 then sends message 3993 to the ISU, which reads the
rest of the modem configuration and specifications from table 3986 at step
3994, and transmits them over the IDL channel at 3995. The HDT stores
pertinent information concerning that ISU for future reference. The
purpose of sending ISU data to the HDT is to accommodate various ISU
models having greatly differing capabilities, and to allow continued use
of legacy ISU equipment when the HDT has been upgraded to include
additional capabilities.
During and after the process of FIGS. 47 and 48, the ISU receiving modem,
FIG. 22 or 23, must track the acquired frequency and symbol timing of the
HDT transmitting modem of FIG. 21. The practicalities of a multicarrier
(MC) system impose requirements which are not obvious from experience with
TDMA (time division multiple-access) and other conventional forms of
bidirectional multipoint networks and systems, nor in point-to-multipoint
("broadcast") multicarrier networks. In TDMA and similar systems, slight
errors in frequency and timing, and larger errors in amplitude, can be
compensated by tracking the received signal. In a broadcast MC system, all
carriers are synchronized at the head end, and can again be tracked at the
receiver. In a bidirectional multipoint-to-point multicarrier system,
however, the HDT receiver must see all channels as though they had been
generated by a single source, because the HDT decodes all channels in an
entire 6 MHz band as a single entity, with a single FFT. Even slight
errors among the individual ISUs in their 10-channel and 130-channel
subbands causes severe distortion and intersymbol interference when the
HDT FFT inverts the channels back into symbol strings for multiple DS0
channels. The errors to be controlled are frequency, symbol timing, and
signal amplitude. Amplitude (power level) in particular has been found to
be much more stringent than in previous systems.
At all times after the ISU receiving modem FIG. 22 or 23 has acquired the
signal of HDT transmitting modem FIG. 21, it must track gradual changes in
frequency, phase, and symbol timing caused by drift and other changes in
the plant. FIG. 49 shows a method 4000 for tracking these changes. Steps
4010, executed by unit 153, FIG. 22 or 155, FIG. 23, track the downstream
power level to compensate for slow gain changes in the plant. Step 4011
measures an average signal level by known methods from the output of FFT
170 or 180. If it is correct, exit 4012 takes no action. If the error is
wrong but within a predetermined threshold of the correct value, block
4013 adjusts a coefficient of equalizer 172 to return the signal level to
its nominal value. If the signal level changes more than a certain amount,
or if it changes rapidly enough, the signal has probably been lost
entirely. In that case, exit 4014 exits tracking mode; it may reenter the
ranging procedure of FIGS. 47 and 48, or it may merely signal an error to
the HDT.
Steps 4020 track carrier frequency in unit 166, FIG. 22 or 23. An HISU
modem of FIG. 23 receives a subband having ten payload channels and one
IOC at the center, as shown in FIG. 16. When this narrowband modem is
tuned to the subband, the sync tones are no longer within its frequency
range. Therefore, step 4021 measures the phase of the IOC channel within
the currently tuned subband, rather than the sync tone used in FIGS. 47
and 48. Step 4022 smooths any phase error between the received carrier and
the locally generated signal from generator 168, FIG. 22, to prevent
jitter. The frame clocks of both the receiving modem FIG. 23 and the
upstream transmitter FIG. 24 use this clock; that is, the clocks are
locked together within the HISU modem. Step 4023 updates the frequency of
a local oscillator in RF synthesizer 157. It should be noted here that the
location of the IOC in the middle of the subband eliminates any offset
phase error which otherwise must be compensated for. Steps 4020 may be the
same for an MISU modem, FIG. 22; this modem has a 130-channel bandwidth,
as shown in FIG. 16. The wider bandwidth of this modem includes multiple
IOC tones for tracking. The modem may use one of them--preferably near the
middle--or pairs of tones offset from the center of the subband.
Step 4030 tracks symbol timing. Step 4031 samples the frequency error
between the received symbols and the local 8 kHz symbol sampling clock. If
the sampling frequencies differ by more than about 2 ppm between the HDT
and the ISU, the synthesized tones progressively depart from their bins at
the receiving FFT until the equalizer can no longer track them. Step 4032
receives the sign of the sampling error from step 4031, and applies a
small 0.5 ppm correction to the frame frequency.
Process 4000 takes places in real time, in parallel with other processes.
After the upstream transmitting modem 101 portion shown in FIG. 24 has
tuned to a subband in FIGS. 47 and 48, it and the upstream receiving modem
82 portion of FIG. 26 must continue to track in amplitude, frequency, and
timing. The use of multicarrier (MC) technology imposes some requirements
which are not obvious from experience with TDMA (time-division multiple
access) or other bidirectional multipoint technologies, nor from
point-to-multipoint (broadcast) MC networks and systems. In TDMA and
similar systems, slight errors in frequency and timing, and larger errors
in signal amplitude, can be compensated by tracking the signal at the
receiver. In broadcast MC systems, all carriers are inherently
synchronized at the head end, and can be tracked adequately at each
receiver separately. In a bidirectional multipoint-to-point multicarrier
plant, however, the head-end receiver sees all channels as though they had
been generated by a single source, because the HDT decodes all channels in
an entire 6 MHz band as a single orthogonal waveform, with a single FFT
converter. Even slight errors among the various ISUs in their 10-channel
and 130-channel subbands can cause severe distortion and intersymbol
interference when the FFT in modem 82 portion of FIG. 26 converts the
channels back into bit strings for multiple DS0 channels. The parameters
to be controlled are frequency, symbol timing, and signal amplitude or
power level. Frequency and timing can be tracked in a manner similar to
steps 4020 and 4030 of FIGS. 47 and 48. Amplitude, however, has been found
to be more critical than in previous systems.
FIG. 50 depicts a method 4100 for tracking changes in the upstream channel
signal amplitude. After FIGS. 47 and 48 have brought the ISU transmitting
modem of FIG. 24 on line, its amplitude must be balanced with that of all
other ISUs in the system. Again, if different upstream subbands were
received by different hardware, or in a TDMA fashion, where amplitude
tracking could be particularized in frequency and/or time, a conventional
AGC circuit could track amplitude variations adequately. In the embodiment
described, power variations greater than about 0.25 dB from one subband to
another cause a significant amount of distortion and intersymbol
interference. In a physical system of, say, a 20 km radius, variations in
the upstream signal level at the head end may vary by 20 dB or more for
different ISU locations, and may additionally vary significantly over time
because of temperature differences, channel loading, component aging, and
many other factors. Conventional methods cannot achieve both the wide
dynamic range and the high resolution required for an MC bidirectional
multipoint-to-point system.
The steps in the left column of FIG. 50 are performed within each ISU; the
HDT performs the steps on the right. Step 4110 selects a number of payload
channels for monitoring from table 4111. The channels must include one
channel from each separate ISU, but need not include more than one. An
MISU thus needs time in only one of its 130 payload channels, a very low
overhead. A 10-channel HISU subband, however, may require time in more
than one of its 10 payload channels, because multiple HISUs can share the
same subband. Of course, a powered-down ISU, or one having no upstream
payload channels provisioned to it, need not participate in blocks 1740,
because it does not transmit upstream at all. (It is alternatively
possible to employ IOC channels instead of payload channels for this
purpose. Although requiring less overhead, such use is generally much more
complex to implement.) After ranging procedure 3900 has acquired a correct
initial power level, step 4120 performs a scan every 30 msec. for all the
selected payload channels, as indicated by arrow 4121. Each ISU responds
at 4130 by sending a message 4131 on its selected upstream channel when
the scan reaches that channel. In step, the HDT measures the received
power level from each ISU separately. If the signal amplitude is within a
certain range of its previous value, then steps 4150 compensate for the
variation at the HDT. Step 4151 smooths the errors over several scans, to
prevent sudden jumps from a single glitch. Step 4152 then adjusts the
coefficients of equalizer 214 in the upstream receiving portion, FIG. 26,
of modem 82, FIG. 3, to compensate for the variation. This sequence
compensates for small, slow variations at a high resolution; the equalizer
steps are small and very linear.
Step 4140 may detect that the HDT equalizer is far from nominal, near the
end of its range--say, 4 dB up or down from nominal, for an equalizer
having a .+-.5 dB range of 0.25 dB steps. This condition might occur for a
large number of accumulated small errors, or it could be caused by a
sudden, major change in the system, such as a break in an optical fiber
followed by an automatic switch to another fiber having a different
length. Steps 4160 compensate for this condition at the transmitting (ISU)
end, rather than at the receiving (HDT) end. The HDT then stops the
channel scan at step 4161, and step 4162 sends an amplitude-error message
4163 over an IOC channel, addressed to the offending ISU. The message
specifies the amount and direction of the compensation to be applied. The
ISU applies this correction at step 4170 to vary the power output of its
transmitter 200, FIG. 24. Conventional transmitter power controls, such as
a PIN attenuator diode 201 in power amplifier 200, are typically
relatively coarse and nonlinear, but they do possess a wide range of
adjustment. DAC 203 receives IOC messages to control attenuator 201. To
allow the head-end equalizer to track the changing power more easily, step
4162 applies the correction over a long period of time, for example, 4-5
sec/dB; but, if the monitored channel is the only channel connected to
that ISU, the entire correction can be made in a single large step.
Instead of controlling adjustment speed at the ISU, the head end may send
individual timed IOC messages for multiple partial corrections; the
downside is increased message traffic on the IOC channels.
ISUs may come online and be powered down at odd times. To prevent a
feckless attempt at correction when an ISU is powered down, or its signal
has been lost for some other reason, step 4140 further detects a condition
of substantially zero power received from the ISU. In that event, step
4180 sets the head-end equalizer to its default value and keeps it there.
Thus, power-leveling blocks 1740 take advantage of the characteristics of
the system to adjust both ends in a way which achieves both high
resolution and wide dynamic range. The digital control available in the
head-end equalizer provides precision and linearity in tracking slow
changes, and the analog control at the ISU provides a wide range, and
still allows the head end to track out inaccuracies caused by its coarse
and nonlinear nature.
Call processing in the communication system 10 entails the manner in which
a subscriber is allocated channels of the system for telephony transport
from the HDT 12 to the ISUs 100. The present communication system in
accordance with the present invention is capable of supporting both call
processing techniques not involving concentration, for example, TR-8
services, and those involving concentration, such as TR-303 services.
Concentration occurs when there are more ISU terminations requiring
service than there are channels to service such ISUs. For example, there
may be 1,000 customer line terminations for the system, with only 240
payload channels which can be allocated to provide service to such
customers.
Where no concentration is required, such as for TR-8 operation, channels
within the 6 MHz spectrum are statically allocated. Therefore, only
reallocation of channels shall be discussed further below with regard to
channel monitoring.
On the other hand, for dynamically allocated channels to provide
concentration, such as for providing TR-303 services, the HDT 12 supports
on-demand allocation of channels for transport of telephony data over the
HFC distribution network 11. Such dynamic allocation of channels is
accomplished utilizing the IOC channels for communication between the HDT
12 and the ISUs 100. Channels are dynamically allocated for calls being
received by a customer at an ISU 100, or for calls originated by a
customer at an ISU 100. The CXMU 56 of HDT 12, as previously discussed,
implements IOC channels which carry the call processing information
between the HDT 12 and the ISUs 100. In particular, the following call
processing messages exist on the IOC channels. They include at least a
line seizure or off-hook message from the ISU to the HDT; line idle or
on-hook message from the ISU to the HDT; enable and disable line idle
detection messages between the HDT and the ISUs.
Call processing in the communication system 10 entails the manner in which
a subscriber is allocated channels of the system for telephony transport
from the HDT 12 to the ISUs 100. The present communication system in
accordance with the present invention is capable of supporting both call
processing techniques not involving concentration, for example, TR-8
services, and those involving concentration, such as TR-303 services.
Concentration occurs when there are more-ISU terminations requiring
service than there are channels to service such ISUs. For example, there
may be 1,000 customer line terminations for the system, with only 240
payload channels which can be allocated to provide service to such
customers.
Where no concentration is required, such as for TR-8 operation, channels
within the 6 MHz spectrum are statically allocated. Therefore, only
reallocation of channels shall be discussed further below with regard to
channel monitoring.
On the other hand, for dynamically allocated channels to provide
concentration, such as for providing TR-303 services, the HDT 12
supports-on-demand allocation of channels for transport of telephony data
over the HFC distribution network 11. Such dynamic allocation of channels
is accomplished utilizing the IOC channels for communication between the
HDT 12 and the ISUs 100. Channels are dynamically allocated for calls
being received by a customer at an ISU 100, or for calls originated by a
customer at an ISU 100. The CXMU 56 of HDT 12, as previously discussed,
implements IOC channels which carry the call processing information
between the HDT 12 and the ISUs 100. In particular, the following call
processing messages exist on the IOC channels. They include at least a
line seizure or off-hook message from the ISU to the HDT; line idle or
on-hook message from the ISU to the HDT; enable and disable line idle
detection messages between the HDT and the ISUs.
For a call to a subscriber on the HFC distribution network 11 the CTSU 54
sends a message to the CXMU 56 associated with the subscriber line
termination and instructs the CXMU 56 to allocate a channel for transport
of the call over the HIFC distribution network 11. The CXMU 56 then
inserts a command on the IOC channel to be received by the ISU 100 to
which the call is intended; the command providing the proper information
to the CXSU 102 to alert the ISU 100 as to the allocated channel.
When a call is originated by a subscriber at the ISU side, each ISU 100 is
responsible for monitoring the channel units for line seizure. When line
seizure is detected, the ISU 100 must communicate this change along with
the PIN address code for the originating line to the CXMU 56 of the HDT 12
using the upstream IOC operation channel. Once the CXMU 56 correctly
receives the line seizure message, the CXMU 56 forwards this indication to
the CTSU 54 which, in turn, provides the necessary information to the
switching network to set up the call. The CTSU 54 checks the availability
of channels and allocates a channel for the call originated at the ISU
100. Once a channel is identified for completing the call from the ISU
100, the CXMU 56 allocates the channel over the downstream IOC channel to
the ISU 100 requesting line seizure. When a subscriber returns on hook, an
appropriate line idle message is sent upstream to the HDT 12 which
provides such information to the CTSU 54 such that the channel can then be
allocated again to support TR-303 services.
Idle channel detection can further be accomplished in the modem utilizing
another technique. After a subscriber at the ISU 100 has terminated use of
a data payload channel, the MCC modem 82 can make a determination that the
previously allocated channel is idle. Idle detection may be performed by
utilizing the equalization process by equalizer 214 (FIG. 26) which
examines the results of the FFT which outputs a complex (I and Q
component) symbol value. An error is calculated, as previously described
herein with respect to equalization, which is used to update the equalizer
coefficients. Typically, when the equalizer has acquired the signal and
valid data is being detected, the error will be small. In the event that
the signal is terminated, the error signal will increase, and this can be
monitored by signal to noise monitor 305 to determine the termination of
the payload data channel used or channel idle status. This information can
then be utilized for allocating idle channels when such operation of the
system supports concentration.
The equalization process can also be utilized to determine whether an
unallocated or allocated channel is being corrupted by ingress as shall be
explained in further detail below with respect to channel monitoring.
The telephony transport system may provide for channel protection from
ingress in several manners. Narrowband ingress is a narrowband signal that
is coupled into the transmission from an external source. The ingress
signal which is located within the OFDM waveform can potentially take the
entire band offline. An ingress signal is (most likely) not orthogonal to
the OFDM carriers, and under worst case conditions can inject interference
into every OFDM carrier signal at a sufficient level to corrupt almost
every DS0+ to an extent that performance is degraded below a minimum bit
error rate.
One method provides a digitally tunable notch filter which includes an
interference sensing algorithm for identifying the ingress location on the
frequency band. Once located, the filtering is updated to provide an
arbitrary filter response to notch the ingress from the OFDM waveform. The
filter would not be part of the basic modem operation but requires the
identification of channels that are degraded in order to "tune" them out.
The amount of channels lost as a result of the filtering would be
determined in response to the bit error rate characteristics in a
frequency region to determine how many channels the ingress actually
corrupted.
Another approach as previously discussed with respect to the ingress filter
and FFT 112 of the MCC upstream receiver architecture of FIG. 26 is the
polyphase filter structure. The cost and power associated with the filter
are absorbed at the HDT 12, while supplying sufficient ingress protection
for the system. Thus, power consumption at the ISUs 100 is not increased.
The preferred filter structure involves two staggered polyphase filters as
previously discussed with respect to FIGS. 31 and 32 although use of one
filter is clearly contemplated with loss of some channels. The
filter/transform pair combines the filter and demodulation process into a
single step. Some of the features provided by polyphase filtering include
the ability to protect the receive band against narrowband ingress and
allow for scalable bandwidth usage in the upstream transmission. With
these approaches, if ingress renders some channels unusable, the HDT 12
can command the ISUs to transmit upstream on a different carrier frequency
to avoid such ingress.
The above approaches for ingress protection, including at least the use of
digital tunable notch filters and polyphase filters, are equally
applicable to point to point systems utilizing multicarrier transport. For
example, a single MISU transporting to a single HDT may use such
techniques. In addition, uni-directional multi-point to point transport
may also utilize such techniques for ingress protection.
In addition, channel monitoring and allocation or reallocation based
thereon may also be used to avoid ingress. External variables can
adversely affect the quality of a given channel. These variables are
numerous, and can range from electro-magnet interference to a physical
break in an optical fiber. A physical break in an optical fiber severs the
communication link and cannot be avoided by switching channels, however, a
channel which is electrically interfered with can be avoided until the
interference is gone. After the interference is gone the channel could be
used again.
Referring to FIG. 40, a channel monitoring method is used to detect and
avoid use of corrupted channels. A channel monitor 296 is used to receive
events from board support software 298 and update a channel quality table
300 in a local database. The monitor 296 also sends messages to a fault
isolator 302 and to channel allocator 304 for allocation or reallocation.
The basic input to the channel monitor is parity errors which are
available from hardware per the DS0+ channels; the DS0+ channels being
10-bit channels with one of the bits having a parity or data integrity bit
inserted in the channel as previously discussed. The parity error
information on a particular channel is used as raw data which is sampled
and integrated over time to arrive at a quality status for that channel.
Parity errors are integrated using two time frames for each of the
different service types including POTS, ISDN, DDS, and DS1, to determine
channel status. The first integration routine is based on a short
integration time of one second for all service types. The second routine,
long integration, is service dependent, as bit error rate requirements for
various services require differing integration times and monitoring
periods as seen in Table 4. These two methods are described below.
Referring to FIGS. 41, 42, and 43, the basic short integration operation is
described. When a parity error of a channel is detected by the CXMU 56, a
parity interrupt is disabled by setting the interrupt priority level above
that of the parity interrupt (FIG. 41). If a modem alarm is received which
indicates a received signal failure, parity errors will be ignored until
the failure condition ends. Thus, some failure conditions will supersede
parity error monitoring. Such alarm conditions may include loss of signal,
modem failure, and loss of synchronization. If a modem alarm is not
active, a parity count table is updated and an error timer event as shown
in FIG. 42 is enabled.
When the error timer event is enabled, the channel monitor 296 enters a
mode wherein parity error registers of the CXMU 56 are read every 10
milliseconds and error counts are summarized after a one second monitoring
period. Generally, the error counts are used to update the channel quality
database and determine which (if any) channels require re-allocation. The
channel quality table 300 of the database contains an ongoing record of
each channel. The table organizes the history of the channels in
categories such as: current ISU assigned to the channel, start of
monitoring, end of monitoring, total error, errors in last day, in last
week and in last 30 days, number of seconds since last error, severe
errors in last day, in last week and in last 30 days, and current service
type, such as ISDN, assigned to the channel.
As indicated in FIG. 41, after the parity interrupt is disabled and no
active alarm exists, the parity counts are updated and the timer event is
enabled. The timer event (FIG. 42), as indicated above, includes a one
second loop where the errors are monitored As shown in FIG. 42, if the one
second loop has not elapsed, the error counts are continued to be updated.
When the second has elapsed, the errors are summarized. If the summarized
errors over the one second period exceed an allowed amount indicating that
an allocated channel is corrupted or bad, as described below, channel
allocator 304 is notified and ISU transmission is reallocated to a
different channel. As shown in FIG. 43, when the reallocation has been
completed, the interrupt priority is lowered below parity so that channel
monitoring continues and the channel quality database is updated
concerning the actions taken. The reallocation task may be accomplished as
a separate task from the error timer task or performed in conjunction with
that task. For example, the reallocator 304 may be part of channel monitor
296.
As shown in FIG. 44 in an alternate embodiment of the error timer task of
FIG. 42, channels can be determined to be bad before the one second has
elapsed. This allows the channels that are determined to be corrupted
during the initial portion of a one second interval to be quickly
identified and reallocated without waiting for the entire one second to
elapse.
Instead of reallocation, the power level for transmission by the ISU may be
increased to overcome the ingress on the channel. However, if the power
level on one channel is increased, the power level of at least one other
channel must be decreased as the overall power level must be kept
substantially constant. If all channels are determined bad, the fault
isolator 302 is notified indicating the probability that a critical
failure is present, such as a fiber break. If the summarized errors over
the one second period do not exceed an allowed amount indicating that the
allocated channel is not corrupted, the interrupt priority is lowered
below parity and the error timer event is disabled. Such event is then
ended and the channels once again are monitored for parity errors per FIG.
41.
Two issues presented by periodic parity monitoring as described above must
be addressed in order to estimate the bit error rate corresponding to the
observed count of parity errors in a monitoring period of one second to
determine if a channel is corrupted. The first is the nature of parity
itself. Accepted practice for data formats using block error detection
assumes that an errored block represents one bit of error, even though the
error actually represents a large number of data bits. Due to the nature
of the data transport system, errors injected into modulated data are
expected to randomize the data. This means that the average errored frame
will consist of four (4) errored data bits (excluding the ninth bit).
Since parity detects only odd bit errors, half of all errored frames are
not detected by parity. Therefore, each parity (frame) error induced by
transport interference represents an average of 8 (data) bits of error.
Second, each monitoring parity error represents 80 frames of data (10
ms/125 .mu.s). Since the parity error is latched, all errors will be
detected, but multiple errors will be detected as one error.
The bit error rate (BER) used as a basis for determining when to reallocate
a channel has been chosen as 10.sup.-3. Therefore, the acceptable number
of parity errors in a one second interval that do not exceed 10.sup.-3
must be determined. To establish the acceptable parity errors, the
probable number of frame errors represented by each observed (monitored)
parity error must be predicted. Given the number of monitored parity
errors, the probable number of frame errors per monitored parity error,
and the number of bit errors represented by a frame (parity) error, a
probable bit error rate can be derived.
A statistical technique is used and the following assumptions are made:
1. Errors have a Poisson distribution, and
2. If the number of monitored parity errors is small (<10) with respect
to the total number of "samples" (100), the monitored parity error rate
(PER) reflects the mean frame error rate (FER).
Since a monitored parity error (MPE) represents 80 frames, assumption 2
implies that the number of frame errors (FEs) "behind" each parity error
is equal to 80 PER. That is, for 100 parity samples at 10 ms per sample,
the mean number of frame errors per parity error is equal to 0.8 times the
count of MPEs in one second. For example, if 3 MPEs are observed in a one
second period, the mean number of FEs for each MPE is 2.4. Multiplying the
desired bit error rate times the sample size and dividing by the bit
errors per frame error yields the equivalent number of frame errors in the
sample. The number of FEs is also equal to the product of the number of
MPEs and the number of FEs per MPE. Given the desired BER, a solution set
for the following equation can be determined.
##EQU1##
The Poisson distribution, as follows, is used to compute the probability of
a given number of FEs represented by a MPE (.chi.), and assumption 2,
above, is used to arrive at the mean number of FEs per MPE (.mu.).
##EQU2##
Since the desired bit rate is a maximum, the Poisson equation is applied
successively with values for .chi. of 0 up to the maximum number. The sum
of these probabilities is the probability that no more than .chi. frame
errors occurred for each monitored parity error.
The results for a bit error rate of 10.sup.-3 and bit errors per frame
error of 1 and 8 are shown in Table 3.
TABLE 3
Bit Error Rate Probability
Maximum
Frame
Errors/
Bit Errors Monitored Monitored Average Frame
per Frame Parity Parity Error Errors/Monitored Probability of
Error Errors (x) Parity Error (.mu.) BER < -10.sup.-3
8 2 4 1.6 98%
3 3 2.4 78%
4 2 3.2 38%
1 8 8 6.4 80%
9 7 7.2 56%
10 7 8.0 45%
Using this technique, a value of 4 monitored parity errors detected during
a one second integration was determined as the threshold to reallocate
service of an ISU to a new channel. This result is arrived at by assuming
a worst case of 8 bit errors per frame error, but a probability of only
38% that the bit error rate is better than 10.sup.-3. The product of the
bit errors per frame, monitored parity errors and maximum frame errors per
monitored parity error must be 64, for a bit error rate of 10.sup.-3 (64
errors in 64 k bits). Therefore, when the sampling of the parity errors in
the error timer event is four or greater, the channel allocator is
notified of a corrupted channel. If the sampled monitored parity errors is
less than 4, the interrupt priority is lowered below parity and the error
timer event is disabled, ending the timer error event and the channels are
then monitored as shown in FIG. 41.
The following is a description of the long integration operation performed
by the background monitor routine (FIG. 45) of the channel monitor 296.
The background monitor routine is used to ensure quality integrity for
channels requiring greater quality than the short integration 10.sup.-3
bit error rate. As the flow diagram shows in FIG. 45, the background
monitor routine operates over a specified time for each service type,
updates the channel quality database table 300, clears the background
count, determines if the integrated errors exceed the allowable limits
determined for each service type, and notifies the channel allocator 304
of bad channels as needed.
In operation, on one second intervals, the background monitor updates the
channel quality database table. Updating the channel quality data table
has two purposes. The first purpose is to adjust the bit error rate and
number of errored seconds data of error free channels to reflect their
increasing quality. The second purpose is to integrate intermittent errors
on monitored channels which are experiencing error levels too low to
result in short integration time reallocation (less than 4 parity
errors/second). Channels in this category have their BER and numbers of
errored seconds data adjusted, and based on the data, may be re-allocated
This is known as long integration time re-allocation, and the default
criteria for long integration time re-allocation for each service type are
shown as follows:
TABLE 4
Service Maximum Errored Monitoring
type: BER: Integration Time: seconds Period:
POTS 10.sup.-3 1 second
ISDN 10.sup.-6 157 seconds 8% 1 hour
DDS 10.sup.-7 157 seconds 0.5% 1 hour
DS1 10.sup.-9 15,625 seconds 0.04% 7 hours
Because POTS service does not require higher quality than 10.sup.-3,
corrupted channels are sufficiently eliminated using the short integration
technique and long integration is not required.
As one example of long integration for a service type, the background
monitor shall be described with reference to a channel used for ISDN
transport. Maximum bit error rate for the channel may be 10.sup.-6, the
number of seconds utilized for integration time is 157, the maximum number
of errored seconds allowable is 8% of the 157 seconds, and the monitoring
period is one hour. Therefore, if the summation of errored seconds is
greater than 8% over the 157 seconds in any one hour monitoring period,
the channel allocator 304 is notified of a bad channel for ISDN transport.
Unallocated or unused channels, but initialized and activated, whether used
for reallocation for non-concentration services such as TR-8 or used for
allocation or reallocation for concentration services such as TR-303, must
also be monitored to insure that they are not bad, thereby reducing the
chance that a bad channel will be allocated or reallocated to an ISU 100.
To monitor unallocated channels, channel monitor 296 uses a backup manager
routine (FIG. 46) to set up unallocated channels in a loop in order to
accumulate error data used to make allocation or re-allocation decisions.
When an unallocated channel experiences errors, it will not be allocated
to an ISU 100 for one hour. After the channel has remained idle
(unallocated) for one hour, the channel monitor places the channel in a
loop back mode to see if the channel has improved. In loop back mode, the
CXMU 56 commands an initialized and activated ISU 100 to transmit a
message on the channel long enough to perform short or long integration on
the parity errors as appropriate. In the loop back mode, it can be
determined whether the previously corrupted channel has improved over time
and the channel quality database is updated accordingly. When not in the
loop back mode, such channels can be powered down.
As described above, the channel quality database includes information to
allow a reallocation or allocation to be made in such a manner that the
channel used for allocation or reallocation is not corrupted. In addition,
the information of the channel quality database can be utilized to rank
the unallocated channels as for quality such that they can be allocated
effectively. For example, a channel may be good enough for POTS and not
good enough for ISDN. Another additional channel may be good enough for
both. The additional channel may be held for ISDN transmission and not
used for POTS. In addition, a particular standby channel of very good
quality may be set aside such that when ingress is considerably high, one
channel is always available to be switched to.
In addition, an estimate of signal to noise ratio can also be determined
for both unallocated and allocated channels utilizing the equalizer 214 of
the MCC modem 82 upstream receiver architecture as shown in FIG. 26. As
described earlier, the equalizer was previously utilized to determine
whether a channel was idle such that it could be allocated. During
operation of the equalizer, as previously described, an error is generated
to update the equalizer coefficients. The magnitude of the error can be
mapped into an estimate of signal to noise ratio (SNR) by signal to noise
monitor 305 (FIG. 26). Likewise, an unused channel should have no signal
in the band. Therefore, by looking at the variance of the detected signal
within the unused FFT bin, an estimate of signal to noise ratio can be
determined. As the signal to noise ratio estimate is directly related to a
probable bit error rate, such probable bit error rate can be utilized for
channel monitoring in order to determine whether a bad or good channel
exists.
Therefore, for reallocation for nonconcentration services such as TR-8
services, reallocation can be performed to unallocated channels with such
unallocated channels monitored through the loopback mode or by SNR
estimation by utilization of the equalizer. Likewise, allocation or
reallocation for concentration services such as TR-303 services can be
made to unallocated channels based upon the quality of such unallocated
channels as determined by the SNR estimation by use of the equalizer.
With respect to channel allocation, a channel allocator routine for channel
allocator 304 examines the channel quality database table to determine
which DS0+ channels to allocate to an ISU 100 for a requested service. The
channel allocator also checks the status of the ISU and channel units to
verify in-service status and proper type for the requested service. The
channel allocator attempts to maintain an optimal distribution of the
bandwidth at the ISUs to permit flexibility for channel reallocation.
Since it is preferred that ISUs 100, at least HISUs, be able to access
only a portion of the RF band at any given time, the channel allocator
must distribute channel usage over the ISUs so as to not overload any one
section of bandwidth and avoid reallocating in-service channels to make
room for additional channels.
The process used by the channel allocator 304 is to allocate equal numbers
of each ISU type to each band of channels of the 6 MHz spectrum If
necessary, in use channels on an ISU can be moved to a new band, if the
current ISU band is fill and a new service is assigned to the ISU.
Likewise, if a channel used by an ISU in one band gets corrupted, the ISU
can be reallocated to a channel in another subband or band of channels.
Remember that the distributed IOC channels continue to allow communication
between the HDT 12 and the HISUs as an HISU always sees one of the IOC
channels distributed throughout the spectrum. Generally, channels with the
longest low-error rate history will be used first. In this way, channels
which have been marked bad and subsequently reallocated for monitoring
purposes will be used last, since their histories will be shorter than
channels which have been operating in a low error condition for a longer
period.
Second Embodiment of Telephony Transport System
A second embodiment of an OFDM telephony transport system, referring to
FIGS. 36-39 shall be described. The 6 MHz spectrum allocation is shown in
FIG. 36. The 6 MHz bandwidth is divided into nine channel bands
corresponding to the nine individual modems 226 (FIG. 37). It will be
recognized by one skilled in the art that less modems could be used by
combining identical operations. Each of the channel bands includes 32
channels modulated with a quadrature 32-ary format (32-QAM) having five
bits per symbol. A single channel is allocated to support transfer of
operations and control data (IOC control data) for communication between
an HDT 12 and ISUs 100. This channel uses BPSK modulation.
The transport architecture shall first be described with respect to
downstream transmission and then with respect to upstream transmission.
Referring to FIG. 37, the MCC modem 82 architecture of the HDT 12 will be
described. In the downstream direction, serial telephony information and
control data is applied from the CXMC 80 through the serial interface 236.
The serial data is demultiplexed by demultiplexer 238 into parallel data
streams. These data streams are submitted to a bank of 32 channel modems
226 which perform symbol mapping and fist Fourier transform (FF)
functions. The 32 channel modems output time domain samples which pass
through a set of mixers 240 that are driven by the synthesizer 230. The
mixers 240 create a set of frequency bands that are orthogonal, and each
band is then filtered through the filter/combiner 228. The aggregate
output of the filter/combiner 228 is then up-converted by synthesizer 242
and mixer 241 to the final transmitter frequency. The signal is then
filtered by filter 232, amplified by amplifier 234, and filtered again by
filter 232 to take off any noise content The signal is then coupled onto
the HFC distribution network via telephony transmitter 14.
On the downstream end of the HFC distribution network 11, an ISU 100
includes a subscriber modem 258 as shown in FIG. 38. The downstream
signals are received from the ODN 18 through the coax leg 30, and are
filtered by filter 260 which provides selectivity for the entire 6 MHz
band. Then the signal is split into two parts. The first part provides
control data and timing information to synchronize clocks for the system.
The second part provides the telephony data. With the control data
received separately from the telephony data, this is referred to as
previously described above as an out of band ISU. The out of band control
channel which is BPSK modulated is split off and mixed to baseband by
mixer 262. The signal is then filtered by filter 263 and passed through an
automatic gain control stage 264 and a Costas loop 266 where carrier phase
is recovered. The signal that results is passed into a timing loop 268 so
timing can be recovered for the entire modem. The IOC control data, which
is a byproduct of the Costas loop, is passed into the 32 channel OFDM
modem 224 of the ISU 100. The second part of the downstream OFDM waveform
is mixed to base band by mixer 270 and associated synthesizer 272. The
output of the mixer 270 is filtered by filter 273 and goes through a gain
control stage 274 to prepare it for reception. It then passes into the 32
channel OFDM modem 224.
Referring to FIG. 39, the IOC control data is hard limited through function
block 276 and provided to microprocessor 225. The OFDM telephony data is
passed through an analog to digital converter 278 and input to a first-in
first-out buffer 280 where it is stored. When a sufficient amount of
information is stored, it is accepted by the microprocessor 225 where the
remainder of the demodulation process, including application of an FFT,
takes place. The microprocessor 225 provides the received data to the rest
of the system through the receive data and receive data clock interface.
The fast Fourier transform (FFT) engine 282 is implemented off the
microprocessor. However, one skilled in the art will recognize that the
FFT 282 could be done by the microprocessor 225.
In the upstream direction, data enters the 32 channel OFDM modem 224
through the transmit data ports and is converted to symbols by the
microprocessor 225. These symbols pass through the FFT engine 282, and the
resulting time domain waveform, including guard samples, goes through a
complex mixer 284. The complex mixer 284 mixes the waveform up in
frequency and the signal is then passed through a random access memory
digital to analog converter 286 (RAM DAC). The RAM DAC contains some RAM
to store up samples before being applied to the analog portion of the ISU
upstream transmitter (FIG. 38). Referring to FIG. 38, the OFDM output for
upstream transport is filtered by filter 288. The waveform then passes
through mixer 290 where it is mixed under control of synthesizer 291 up to
the transmit frequency. The signal is then passed through a processor gain
control 292 so that amplitude leveling can take place in the upstream
path. The upstream signal is finally passed through a 6 MHz filter 294 as
a final selectivity before upstream transmission on the coaxial leg 30 to
the ODN 18.
In the upstream direction at the HDT 12 side, a signal received on the coax
from the telephony receiver 16 is filtered by filter 244 and amplified by
amplifier 246. The received signal, which is orthogonally frequency
division multiplexed, is mixed to baseband by bank of mixers 248 and
associated synthesizer 250. Each output of the mixers 248 is then filtered
by baseband filter bank 252 and each output time domain waveform is sent
then to a demodulator of the 32 channel OFDM modems 226. The signals pass
through a FFT and the symbols are mapped back into bits. The bits are then
multiplexed together by multiplexer 254 and applied to CXMC 56 through the
other serial interface 256.
As shown in this embodiment, the ISU is an out of band ISU as utilization
of separate receivers for control data and telephony data is indicative
thereof as previously discussed. In addition, the separation of the
spectrum into channel bands is further shown Various other embodiments as
contemplated by the accompanying claims of the transport system are
possible by building on the embodiments described herein. In one
embodiment, an IOC control channel for at least synchronization
information transport, and the telephony service channels or paths are
provided into a single format. The IOC link between the HDT 12 and the
ISUs 100 may be implemented as four BPSK modulated carriers operating at
16 kbps, yielding a data rate of 64 kbps in total. Each subscriber would
implement a simple separate transceiver, like in the second embodiment,
which continuously monitors the service channel assigned to it on the
downstream link separately from the telephony channels. This transceiver
would require a tuned oscillator to tune to the service IOC channel.
Likewise, an IOC channel could be provided for channel bands of the 6 MHz
bandwidth and the channel bands may include orthogonal carriers for
telephony data and an IOC channel that is received separately from the
reception of the orthogonal carriers.
In another embodiment, instead of 4 BPSK channels, a single 64 kbps IOC
channel is provided. This single channel lies on the OFDM frequency
structure, although the symbol rate is not compatible with the telephony
symbol rate of OFDM framework. This single wide band signal requires a
wider band receiver at the ISU 100 such that the IOC link between the HDT
12 and ISUs is always possible. With single channel support it is possible
to use a fixed reference oscillator that does not have to tune across any
part of the band in the subscriber units. However, unlike in the first
embodiment where the IOC channels are distributed across the spectrum
allowing for narrow band receivers, the power requirements for this
embodiment would increase because of the use of the wide band receiver at
the ISU 100.
In yet another embodiment, the IOC link may include two IOC channels in
each of 32 OFDM channel groups. This increases the number of OFDM carriers
to 34 from 32 in each group. Each channel group would consist of 34 OFDM
channels and a channel band may contain 8 to 10 channels groups. This
approach allows a narrow band receiver to be used to lock to the reference
parameters provided by the HDT 12 to utilize an OFDM waveform, but adds
the complexity of also having to provide the control or service
information in the OFDM data path format. Because the subscriber could
tune to any one of the channel groups, the information that is embedded in
the extra carriers must also be tracked by the central office. Since the
system needs to support a timing acquisition requirement, this embodiment
may also require that a synchronization signal be place off the end of the
OFDM waveform.
It is to be understood, however, that even though numerous characteristics
of the present invention have been set forth in the foregoing description,
together with details of the structure and function of the invention, the
disclosure is illustrative and changes in matters of order, shape, size,
and arrangement of the parts, and various properties of the operation may
be made within the principles of the invention and to the full extent
indicated by the broad general meaning of the terms in which the appended
claims are expressed.
In the following detailed description of the preferred embodiments,
reference is made to the accompanying drawings that form a part hereof,
and in which are shown by way of illustration specific embodiments in
which this invention may be practiced. It is understood that other
embodiments may be used and structural changes may be made without
departing from the scope of the claimed invention.
FIG. 70 shows one embodiment of this invention having an apparatus,
generally indicated as FFT system 2100, which performs both forward and
inverse FFT functions. The input/output signals to FFT system 2100 include
some (or, in one preferred embodiment, all) of the following: real data-in
2111 (having an N-bit-wide data path; in one embodiment, this is 10 bits
wide, and bi-directional so it can be both written to and read from),
imaginary data-in 2112 (having an N-bit-wide data path; in one embodiment,
this is 10 bits wide, and bi-directional so it can be both written to and
read from), address in 2113 (having enough bits to specify addresses for
each input value or input-value pair), control and clock lines 2114 that
control writing (and reading) data from the input side, test signals 2115,
size-select bits 2116, bit-growth-select bits for each of P stages
including bit-growth-1 bits 2117 through bit-growth-P bits 2118,
forward/inverse select signal 2119 which specifies whether to perform a
forward or inverse transform, power-down command signal input 2109, real
data-out 2121 (having an M-bit-wide data path, in one embodiment, this is
10 bits), imaginary data-out 2122 (having an M-bit-wide data path, in one
embodiment, this is 10 bits), address out 2123 (having enough bits to
specify addresses for each output value or output value pair)(in one
embodiment, these are driven by an external device to select output values
as they are needed), control and clock lines 2124 that control reading
data from the output side, overflow signal 2125 which indicates that one
or more output values has overflowed, and FFT complete signal 2126 which
is activated when a transform has been completed.
ASIC 2101 Fabrication
In one embodiment, the FFT system 2100 is fabricated on an
application-specific integrated circuit ("ASIC") 2101, a chip fabricated
by LSI Logic Inc. In this embodiment, full scan testing circuits are
included into the ASIC 2101 for testability. In this embodiment, FFT
system 2100 is fabricated in LSI Logic LCB500K technology, which is a 0.5
micron rule, 3.3 Volt CMOS (complementary metal-oxide semiconductor)
process.
Functional Description, Overview of FFT System 2100
In one embodiment, ASIC 2101 has four pins, size select 2116, to select
between the various transform sizes (i.e., transforms having 2.sup.N
points; where 5.ltoreq.N.ltoreq.10, thus providing selectability for a
1024-point transform, a 512-point transform, a 256-point transform, a
128-point transform, a 64-point transform, or a 32-point transform). In
one application, transforms are completed in less than 125 microseconds.
See the "Execution Time" section below for the minimum clock frequencies
necessary to meet this requirement. In one embodiment, when performing a
1024-point transform, a clock of at least approximately 32 MHz is
required. In one embodiment, when performing a 512-point transform, a
clock of at least approximately 16.5 MHz is required. In one embodiment,
when performing a 32-point transform, a clock of at least approximately 4
MHz is required.
FIG. 71 is a block diagram of modem 2400 according to the present invention
which includes a FFT system 2100 configured to perform an IFFT in
transmitter section 2401 (similar to the transmitter shown in FIG. 21) and
another FFT system 2100 configured to perform an FFT in receiver section
2402 (similar to the receiver shown in FIG. 26).
In one embodiment, ASIC 2101 has three logical banks of RAM which are
configurable as shown in FIG. 72: an input RAM 2251 (containing a
plurality of real input values 2241 and a plurality of imaginary input
values 2245), an output RAM 2253 (containing a plurality of real output
values 2243 and a plurality of imaginary output values 2247), and a
conversion RAM 2252 (containing a plurality of real conversion values 2242
and 2244 and a plurality of imaginary conversion values 2246 and 2248). In
one embodiment, input RAM 2251 has 1024 complex-value positions, each 20
bits wide (10 bits wide for each real input value 2241 and 10 bits wide
for each imaginary input value 2245), output RAM 2253 has 1024
complex-value positions, each 20 bits wide (10 bits wide for each real
output value 2243 and 10 bits for each imaginary output value 2247) and
conversion RAM 2252 has 1024 complex-value positions, each 30 bits wide
(15 bits wide for each real conversion value 2242 and 2244 and 15 bits
wide for each imaginary conversion value 2246 and 2248). (The 5 extra
low-order real bits 2244 and 5 extra low-order imaginary bits 2248 in each
position of the conversion RAM 2252 help to avoid loss of precision during
calculations.) The internal functions of ASIC 2101 have exclusive access
to the conversion RAM 2252 and perform the FFT calculations out of this
conversion RAM 2252. The input RAM 2251 is accessible to the user's input
device as an input to the FFI system 2100 (to be written under external
control with input data). The output RAM 2253 is accessible to the user's
output device as an output source (to be read under external control to
obtain output data). In one embodiment, the rising edge of an
approximately 8-KHz frame clock 2108 is used to start the computation of
each transform (e.g., either a 1024-point FFT, or a 1024-point IFFT).
In one embodiment, the functions of all three banks of RAMs are also
"permuted" on the rising edge of this 8-KHz frame clock 2108, before each
FFT calculation starts. As used herein, this "permutation" changes the
function of each bank of RAM without actually moving data: input RAM 2251
becomes conversion RAM 2252, conversion RAM 2252 becomes output RAM 2253,
and output RAM 2253 becomes input RAM 2251. Note that, in this embodiment,
no RAM data is moved when the functions are permuted. FIG. 73 shows one
embodiment of a physical implementation which provides the function of
input RAM 2251, conversion RAM 2252, and output RAM 2253. Bank control
block 2131 permutes the function of the physical RAM banks 2151, 2152 and
2153 at the rising edge of frame clock 2139. One of the functions of bank
control state machine 2131 is to control the routing of data through RAM
input multiplexers (MUXs) 2171, 2172, and 2173 and the routing of data
through RAM output multiplexers (MUXs) 2181, 2182, and 2183. For example,
at a first given state, bank control 2131 controls input-select block 2132
to input data into physical RAM bank 2151 (in one embodiment, physical RAM
bank 2151 includes 1024 ten-bit real values and 1024 ten-bit imaginary
values). Thus, in the first state, physical RAM bank 2151 acts as logical
input RAM bank 2251. Once all of the desired first set of input values (up
to 1024 values or pairs of values) have been inputted, the frame clock
2139 is driven to change the state of bank control 2131 (permuting the
three RAM bank's respective functions) to a second state, in which bank
control 2131 controls input-select block 2132 to input data into physical
RAM bank 2152 and bank control 2131 controls conversion-select block 2133
to direct computation accesses for data into physical RAM bank 2151. (The
five low-order bits of values used for computation are always provided
from physical RAM bank 2154.) Thus, in the second state, physical RAM bank
2152 acts as logical input RAM bank 2251 and physical RAM bank 2151 acts
as logical conversion RAM bank 2252. Once all of the desired second set of
input values (up to 1024 values or pairs of values) have been inputted,
the frame clock 2139 is driven to change the state of bank control 2131
(permuting the three RAM bank's respective functions again) to a third
state, in which bank control 2131 controls input-select block 2132 to
input data into physical RAM bank 2153, bank control 2131 controls
conversion-select block 2133 to direct computation accesses for data into
physical RAM bank 2152, and bank control 2131 controls output-select block
2134 to direct output requests for accesses for data from physical RAM
bank 2151. Thus, in the third state, physical RAM bank 2153 acts as
logical input RAM bank 2251, physical RAM bank 2152 acts as logical
conversion RAM bank 2252, and physical RAM bank 2151 acts as logical
output RAM bank 2253. Once all of the desired third set of input values
(up to 1024 values or pairs of values) have been inputted--and the first
set of converted output values (up to 1024 values or pairs of values) have
been outputted--then the frame clock 2139 is driven to change the state of
bank control 2131 (permuting the three RAM bank's respective functions
again) back to the first state, in which bank control 2131 controls
input-select block 2132 to input data into physical RAM bank 2151, bank
control 2131 controls conversion-select block 2133 to direct computation
accesses for data into physical RAM bank 2152, and bank control 2131
controls output-select block 2134 to direct output requests for accesses
for data from physical RAM bank 2153. Thus, in the first state, physical
RAM bank 2151 again acts as logical input RAM bank 2251, physical RAM bank
2153 acts as logical conversion RAM bank 2252, and physical RAM bank 2152
acts as logical output RAM bank 2253.
At the beginning of each transform (i.e, FFT/IFFT (fast Fourier transform
or inverse fast Fourier transform)) process, the functions of the RAM bank
memories (the mapping of RAMs 2151, 2152, and 2153 to RAMs 2251, 2252, and
2253) are permuted. The conversion RAM 2252 becomes the output RAM 2253,
the input RAM 2251 becomes the conversion RAM 2252, and the output RAM
2253 becomes the input RAM 2251. Each RAM (2251,2252, and 2253) has its
own independent control and clock signals (2114, 2128, and 2124,
respectively). While the current transform is being calculated in
conversion RAM 2252, the results of the previously calculated FFT/IFFT may
be read from the output RAM 2253 and the data for the next FFT calculation
may be asynchronously and simultaneously written to the input RAM 2251. In
one embodiment, access to the input RAM 2251 and output RAM 2253 is
restricted for 3 clock cycles (one prior to and two after) relative to the
rising edge of the frame clock. This allows the RAM permutation to proceed
safely without undesirable data loss.
Forward Fast Fourier Transform (FFT) and
Inverse Fast Fourier Transform (IFFT)
In one embodiment, FFT system 2100 is implemented on a single integrated
circuit (IC) that performs both a forward FFT and an inverse FFT ("IFFT",
also called a reverse FFT). A pin, forward/inverse signal 2119, selects
between the two types of transform. The inverse FFT uses the identical
calculation sequence as the forward transform, but the complex values
(i.e., real and imaginary) of the twiddle factors and butterfly
coefficients are conjugated relative to these values used for the FFT. A
forward FFT is defined to convert a time-domain signal into
frequency-domain signals and, in one embodiment, is used in the
transmitter 2401 of modem 2400 (see FIG. 71). An inverse FFT is defined to
convert frequency-domain signals to time-domain signals and is used in the
receiver 2402 of the modem 2400.
In one embodiment, the underlying structure of FFT system 2100 supports
five radix-4 butterflies (i.e., butterfly operations) usable to perform
the 1024-point transforms (FFT and IFFT). The butterflies and stages are
reduced for lower-order transforms. (In one embodiment, every other
twiddle factor of the 1024-point twiddle-factor lookup table 2610 is used
when calculating the 512-point transforms by forcing to zero the low-order
address bit to the twiddle-factor lookup table 2610. Every fourth twiddle
factor of the 1024-point twiddle-factor lookup table 2610 is used when
calculating the 256-point transforms by forcing to zero the two low-order
address bits to the twiddle-factor lookup table 2610.) In one embodiment,
all FFT transforms are calculated by using radix-4 butterflies, except for
the last stage of the 512-point, 128-point and 32-point transforms, which
use the radix-4 structure to perform a radix-2 butterfly. (Obvious
extensions are made if other transforms are used). Thus, the 1024-point
transforms use five stages of radix-4 butterflies, the 512-point
transforms use four stages of radix-4 butterflies followed by one stage of
a radix-2 butterfly, etc.
Scaling of FFT output
In one embodiment, scaling is controlled by ten external pins (bit-growth
signals 2117 though 2118) on the ASIC 2101. Two pins are used for each of
the five passes required for the 1024- and 512-point transform. The use of
two pins at each stage allows a scaling factor (e.g., a shift right of
each intermediate result value) of from 0 bits to 3 bits which exceeds the
nominal bit growth of 1 bit observed in the inventor's analysis. In one
embodiment, the binary number represented by each pair of pins indicates
the number of binary places that the results of each calculation are to be
shifted right (divided) by before they are placed back into conversion RAM
2252. Note that in one such embodiment scaling the result after each
butterfly operation requires at least some working registers to maintain
bits greater than the MSB of the 15-bit values (each for real and
imaginary) in the internal conversion RAM 2252, and the scaling to be
applied prior to storage in the conversion RAM 2252.
Round-off/truncation, saturation, and scaling
In one embodiment, each input and output number is represented by 10 bits
(i.e., 10 bits for each real portion, plus 10 bits for each imaginary
portion of a complex pair of numbers). Numbers are represented as
two's-complement fractional arithmetic with the binary point immediately
to the right of the sign bit. Fractional arithmetic helps prevent
multiplies from causing overflow. The result portion of a double-precision
multiply which is used is the upper fifteen bits.
Number growth is managed by the 10 scaling pins (bit growth signals
2117-2118) mentioned previously. In addition, the ASIC 2101 uses
saturation logic that prevents number roll over. That is to say, that if
the result of an addition or substraction exceeds the maximum value that
can be represented in 15 bits, the result is replaced by a value
representing the maximum possible integer. Likewise, if the result of an
addition or substraction is less than the minimum value that can be
represented in 15 bits, the result is replaced by a value representing the
minimum possible integer.
In one embodiment, ASIC 2101 has the following registers for real and
complex data: 10-bit input RAM 2251 and 10-bit output RAM 2253, and a 15
bit conversion RAM 2252. The input values are presented as a two's
complement fractional binary value (2.bbbbbbbbb). In the internal
conversion RAM, the five extra bits are appended at the bottom (i.e.,
low-order position) of the input word. The resulting format is an extended
2's-complement value in the internal conversion RAM 2252 (e.g.,
s.bbbbbbbbbxxxxx, where s is sign, b's are significant input bits, and x's
are extra bits to maintain precision). Note that in one embodiment any
extension bits in the working registers extend the sign bit to achieve the
desired 2's-complement results.
In one embodiment, at the output of each transform pass, the result is
shifted down as specified by the signals provided into the scaling pins
2117-2118 for that pass. The output RAM 2253 aligns precisely with the
upper 10 bits of the conversion RAM 2252, in a manner similar to the input
RAM 2251 (s.bbbbbbbbbxxxxx), each output value (after the final growth
factor is applied prior to storage in the internal conversion RAM 2252 is
rounded to produce the output results (s.bbbbbbbbr).
Power Down
In one embodiment, the ASIC has a single pin, power-down signal 2109, to
control power down that gates off the internal clock in the ASIC, and
places the input RAM 2251 and output RAM 2253 into a low power state. In
one embodiment, access attempts to those RAMS 2251 and 2253 while in the
low power state will be unsuccessful.
Execution Time
The maximum FFT execution frequency requires that a 1024-point transform be
completed in 125 microseconds based on a 32 MHz clock.
An approximately 8-KHz frame clock 2108 is provided to the ASIC 2101 to
signify the beginning of a transform cycle. Prior to the rising edge of
this clock, the previous FFT should be complete. Table 5 below summarizes
the clock frequency to perform a 1024-point and a 512-point FFT in 125
microseconds.
TABLE 5
FFT Clock required to perform a transform in 125 microseconds
Clock Frequency for 125
Clock cycles per .mu.Sec conversion rate
Transform Size conversion (MHz)
1024 3845 30.8
512 1925 15.4
256 775 6.2
128 390 3.1
64 150 1.2
32 80 0.64
FFT Functional Blocks
The ASIC is partitioned into the following major functional blocks.
1. RAM banks 2151, 2152, 2153
2. Sequencer 2640
3. Dual radix (2, 4) core 2600, including multipliers 2620 through 2627
4. Twiddle-factor lookup table 2610
The algorithm used is a decimation in time (DIT) FFT. The algorithm used is
an in place algorithm which means that the results of each butterfly are
put back into the same locations that the operands came from. The
algorithm assumes digit reversed input order and normal output order.
Since the three banks of RAMs are independent, the reordering of the input
data is done internally to the ASIC. This reordering is transparent to the
user. Thus the user writes the data into the input RAM bank in normal
order and read the data from the output RAM bank in normal order. If the
user requires some other ordering, this may be accomplished by simply
permuting the address line ordering.
The order of the memory for all transform sizes are given in the section
titled "Sequencer."
Memory Banks
The ASIC 2101 includes three banks of RAM that are individually
addressable. The RAM banks 2251, 2253, and 2252 are used for input data,
output data and conversion data, respectively. The three banks are used in
order to obtain real-time execution of the FFT and to allow independent
clock rates for the input and output of data in order to accommodate
clocking needs of those devices connected to it. In one embodiment, the
input RAM bank 2251 and output RAM bank 2252 are each organized as 1024
twenty-bit words. The lower 10 bits of each 20-bit word are used for real
data, and the upper 10 bits of each 20-bit word are used for imaginary
data The conversion RAM bank 2252 is organized as 1024 thirty-bit words.
In one embodiment, the lower 15 bits of each thirty-bit word are used for
real data and the upper 15 bits for imaginary data. The connectivities of
the three RAM banks of memory are controlled by a state machine which is
advanced by the 8-KHz frame signal 2108. The state machine has three
states which are defined in Table 6 as follows:
TABLE 6
Typical processing states of memory banks.
State BANK A BANK B BANK C
0 Input Convert Output
1 Convert Output Input
2 Output Input Convert
In one embodiment, the RAM banks do not have a reset. For input and output,
they are accessed at a maximum of 10.24 MHz rate. In one embodiment, a
dead time of one clock cycle before the rising edge of the 8-KHz clock
2108 and two clock cycles after is required to assure the safety of the
RAM bank switching. The ASIC 2101 clocks the data in on the rising edge of
the input clock and clocks data out on the rising edge of the output
clock.
Actual operation of the conversion RAM 2252 is a bit more complicated than
is implied above. Actually only the top 10 bits of the conversion RAM 2252
participates in the bank switching between RAMs 2151, 2152, and 2153. The
lower 5 bits 2154 are dedicated the conversion RAM 2252. Since all numbers
are MSB aligned, no shifting needs to be done on input or output. However,
the last stage of the FFT calculation rounds the results to ten bits. This
eliminates a bias that would result had the results simply been truncated.
The rounding is accomplished by adding a one to the eleventh bit position
and then truncating this result.
Sequencer 2640
The sequencer 2640 manages the processing of the FFT system 2100. The
sequencer 2640 controls the generation of addresses for the conversion RAM
bank 2252 and the twiddle factor ROMs 2610 through address generation
blocks 2642 and 2641 respectively. Sequencer 2640 also configures the
calculation commands for the radix butterfly calculator 2630. In addition,
sequencer 2640 monitors the calculations for overflow. If at any time
during the course of FFT calculation, an overflow or underflow is
detected, then a flag is set indicating that the results of the FFT are
suspect. This overflow flag is passed along with the output data block
when the RAM banks are switched. Thus the flag pin 2125 indicates that the
output data block presently being read out of the output RAM bank 2253 may
not be accurate. This flag 2125 may aid the system designer in providing
an indicator for AGC (automatic gain control).
Dual radix (2, 4) core 2600
The dual radix core 2600 is the arithmetic element of the ASIC 2101. In one
preferred embodiment, it includes eight 16-bit-by-16-bit multipliers (2620
through 2627) and thirty-two multiplexed adder-subtractor-accumulators
2633. In another embodiment, it includes twelve 16-bit-by-16-bit
multipliers and eight multibranched adder/subtractor/accumulators.
FIG. 74 shows one embodiment of a dual radix core 2600. In the embodiment
shown in FIG. 74, conversion RAM bank 2252 is shown as part of the dual
radix core. In other embodiments, conversion RAM bank 2252 is a separate
functional unit, not considered part of the dual radix core 2600. In FIG.
74, the data fetched from conversion RAM bank 2252 is fetched into holding
latches 2612, with both the real (i.e., X3R, X2R, X1R, and X0R) and
imaginary parts (i.e., X3I, X2I, X1I, and X0I) of four points being
fetched in parallel, substantially simultaneously. As shown in FIG. 74,
the real and imaginary data fetched from conversion RAM bank 2252 is
addressed by address generation circuit 2642 which provides four separate
addresses to conversion RAM bank 2252. The real and imaginary part of a
single point both use the same address. The four separate addresses allow
different sets of points to be fetched simultaneously during various
stages of the transform operation.
As further shown in FIG. 74, the data fetched from conversion RAM bank 2252
includes real-and-imaginary pairs denoted X3, X2, X1 and X0. The data
point pairs being denoted are X3 which includes X3R and X3I, X2 which
includes X2R and X2I, X1 which includes X1R and X1I, and X0 which includes
X0R and X0I. In one embodiment, the data fetched from conversion RAM bank
2252 is held for two successive multiplier clock cycles in holding latches
2612, so that the same value gets multiplied in each of two multiply
cycles. The corresponding twiddle factors are fetched on every clock cycle
from twiddle factor look-up table 2610. For example in one embodiment, on
even clock cycles the real components for four twiddle factors are fetched
in parallel denoted W0R, W1R, W2R, and W3R. These four real twiddle
factors are fed to multipliers 2620 and 2621, 2622 and 2623, 2624 and
2625, and 2626 and 2627, respectively as shown, and multiplied by the real
and imaginary components of all four data value pairs fetched from
conversion RAM bank 2252. The products of these eight multipliers
2620-2627 are fed through routing logic 2634 into all thirty-two
adder/subtractor accumulators 2633 of row-column array 2632. On the
successive odd multiplier clock cycle, four imaginary twiddle factors are
then fetched, shown as W0I, W1I, W2I and W3I. As shown, imaginary twiddle
factor W0I is fed to both multiplier 2620 and multiplier 2621. Similarly,
imaginary twiddle factor W1I is fed to both multiplier 2622 and multiplier
2623. Imaginary twiddle factor W2I is fed to multiplier 2624 and 2625 and
imaginary twiddle factor W3I is fed to multiplier 2626 and 2627. For the
second multiplier clock, these eight multipliers are again fed with the
same complex point data still held in holding latches 2612 as was used in
the first clock multiplier cycle. Again after the second clock multiplier,
the eight products are fed through routing logic 2634 and accumulated in
adder/subtractor accumulators 2633. Note that each twiddle factor value in
twiddle factor lookup table 2610 is a complex number having a real and
imaginary part (in this embodiment, the real and imaginary parts are
stored in successive locations which are fetched on successive clock
cycles, thus presenting 4 real values followed by four imaginary values)
and each point value in conversion RAM bank 2252 is also a complex number
having a real part and an imaginary part (in this embodiment, the real and
imaginary parts are stored in side-by-side locations which are fetched on
only every other clock cycle, thus presenting 4 real values and four
imaginary values on every other clock cycle).
These elements are configured to perform a radix-4 butterfly calculation.
The radix-4 butterfly calculations needed to provide a 1024-point FFT or
IFFT and the selection of addresses and twiddle coefficients for the,
e.g., 1024 points used for each of the five passes needed for a complete
1024-point transform are easily calculated and are well known in the art
(see, for example, chapters 6 & 10 of Theory and Application of Digital
Signal Processing, by Lawrence R. Rabiner and Bernard Gold, Prentice-Hall
Inc, published in 1975; and page 612 et seq. of Discrete-Time Signal
Processing, by Alan V. Oppenheim and Ronald W. Schafer, Prentice-Hall Inc,
published in 1989). Although a radix-2 transform is also required, a
separate radix-2 calculator is not required because the radix-2 operation
is a subset of the radix-4 operation. Only the operands actually needed
for the radix-2 operation are actually loaded. The remainder are set to
zero. All arithmetic is performed in such a manner that, if an overflow or
underflow should occur, then the results saturate. This prevents roll-over
from contaminating the results. The function performed by the dual-radix
core 2600 for a radix-4 butterfly operation supporting the forward
transform is the following:
A'=A+BW.sup.1k +CW.sup.2k +DW.sup.3k
B'=A-jBW.sup.1k -CW.sup.2k +jDW.sup.3k
C'=A-BW.sup.1k +CW.sup.2k -DW.sup.3k
D'=A+jBW.sup.1k -CW.sup.2k -jDW.sup.3k
where A, B, C and D are the four input points, A', B', C' and D' are the
four output points, and W.sup.1k, W.sup.2k and W.sup.3k are the twiddle
coefficients.
The function performed in a reverse transform is simply the complex
conjugate of the above set of equations.
A'=A+BW.sup.1k +CW.sup.2k +DW.sup.3k
B'=A+jBW.sup.1k -CW.sup.2k -jDW.sup.3k
C'=A-BW.sup.1k +CW.sup.2k -DW.sup.3k
D'=A-jBW.sup.1k -CW.sup.2k +jDW.sup.3k.
Note that the twiddle-factor W which is used for each A on the right side
of the above equations is one (i.e., the complex number 1+j0).
The multiplication of two complex numbers, each having a real part and an
imaginary part, for example results in the following equation:
##EQU3##
Thus four multiplier operations are needed for each complex multiply
operation.
In order to speed the transform function, the factors for the parallel
multipliers are fetched in parallel under the control of control and
clocking sequencer 2640. Routing logic 2634 routes the products of the
multipliers 2620-2627 to the thirty-two adder-subtractor-accumulators
2633. In one embodiment, eight multiplier cycles, C0 through C7, are used
to generate four radix-4 butterfly operations, resulting in sixteen
complex output values. For discussion purposes, row-column array 2632 is
shown having four rows (A, B, C, and D) and four columns (W, X, Y, and Z)
of complex value pairs. The real value and the imaginary value of each of
these sixteen complex value pairs has its own associated
adder-subtractor-accumulator 2633, for a total of thirty-two
adder-subtractor-accumulators 2633, as shown in FIG. 74 and FIG. 91. In
one embodiment, scaling-factor shift logic 2644 (under the control of
bit-growth selector 2643) is provided in the path between
adder-subtractor-accumulators 2633 and conversion RAM bank 2252. The
scaling-factor shift logic 2644 provides a right-shift function of 0 bits,
1 bit, 2 bits or three bits (divide by 1, two, four or eight respectively)
on each output data value as it is being returned to conversion RAM 2252.
Bit-growth pins 2117 through 2118, which control the divide function for
each of the passes are coupled to bit-growth selector 2643 under control
of sequencer 2640.
FIGS. 75-82 are a table 2800 showing the order of calculations for a
"normal butterfly sub-operation." The data points in conversion RAM 2252
are arranged within conversion RAM 2252 such that the four input points
for one radix-4 operation are each located in different sub-banks if the
points are successively addressed (e.g., addresses 0, 1, 2, and 3 are each
in different sub-banks, e.g., sub-banks 2290, 2291, 2292, and 2293
respectively), but points whose addresses differ by a factor of 4 are
located in the same bank (e.g., addresses 0, 4, 8, and 12 are all within
bank 2290, as are addresses 0, 16, 32, and 48, addresses 0, 64, 128, and
192, and addresses 0, 256, 512, and 768). The butterfly passes for this
second set of points (those whose addresses-mod-4 are equal) are handled
by the equations shown in the table of FIGS. 75-82. FIG. 75 shows the
operations at each of the thirty-two adder-subtractor-accumulators 2633 at
a multiplier clock cycle command denoted C0. For example, at C0, the
adder-subtractor-accumulator 2633 for the real portion of the AW point in
row-column array 2632 (called the AWR accumulator) gets loaded with the
output (called WR) of multiplier 2620, and the
adder-subtractor-accumulator 2633 for the imaginary portion of the DZ
point in row-column array 2632 (called the DZI accumulator) gets loaded
with the output (called ZI) of multiplier 2627. By performing load
operations at clock C0, the previous values of the accumulators do not
need to be zeroed. As shown in FIG. 91, multipliers 2620, 2621, 2622,
2623, 2624, 2625, 2626 and 2627 produce products called WR, WI, XR, XI,
YR, YI, ZR and ZI, respectively, however the -R and -I designations of
these products are not strictly correlated to real and imaginary numbers.
FIG. 91 also shows the row and column locations for the thirty-two
adder-subtractor-accumulators 2633, with AWR shown in the upper-left
corner and DZI in the lower right corner.
FIG. 76 shows the operations at each of the thirty-two
adder-subtractor-accumulators 2633 at a multiplier clock cycle command
denoted C1. For example, at C1, the adder-subtractor-accumulator 2633 for
the real portion of the AW point in row-column array 2632 (called the AWR
accumulator) gets loaded with the difference of subtracting from its
previous value (called AWR, this value happens to be the WR value loaded
in cycle CO) the output (called WI) of multiplier 2621, and the
adder-subtractor-accumulator 2633 for the imaginary portion of the DZI
point in row-column array 2632 (called the DZI accumulator) gets loaded
with the sum of its previous value (called DZI, this value happens to be
the ZI value loaded in cycle C0) and output (called ZR) of multiplier
2626.
Similarly, FIGS. 77 through 82 show the operations which take place at
multiplier clocks C2 through C7, respectively.
Since each complex-multiply operation takes a total of four multiplier
operations, and two multipliers (e.g., the pair 2620 and 2621) are used,
two multiplier cycles are needed for each complex-multiply operation. In a
1024-point transform (i.e., either an FFT or an IFFT), four of the five
passes involve sets of four points wherein all four points are in a single
sub-bank (e.g., 2290), and therefore must be fetched on four successive
even-clocks. Each of these four passes takes eight clocks, called C0
through C7. These four passes are each called "normal butterfly." Table
2800 shows the order of calculation for all of the suboperations for one
embodiment of a normal butterfly (calculating four radix-4 butterfly
operations in eight multiplier clock cycles), where each of the four
points for one radix-4 butterfly are in the same sub-bank (e.g., either
sub-bank 2290 or 2291 or 2292 or 2293).
FIGS. 83-90 are a table 2810 showing the order of calculations for a
"transposed butterfly sub-operation." The transposed butterfly
sub-operation is used for one pass of each transform performed. The data
points in conversion RAM 2252 are arranged within conversion RAM 2252 such
that the four input points for one radix-4 operation are each located in
different sub-banks if the points are successively addressed (e.g.,
addresses 0, 1, 2, and 3 are each in different sub-banks, e.g., sub-banks
2290, 2291, 2292, and 2293 respectively). The transposed butterfly passes
for this one set of points (those whose addresses-mod-4 are equal) are
handled by the equations shown in the table of FIGS. 83-90. FIG. 83 shows
the operations at each of the thirty-two adder-subtractor-accumulators
2633 at a multiplier clock cycle command denoted C0 (note that only eight
adder-subtractor-accumulators 2633 are affected, the other twenty-four do
nothing). For example, at C0, the adder-subtractor-accumulator 2633 for
the real portion of the AW point in row-column array 2632 (called the AWR
accumulator) gets loaded with the result of a four-way addition of the
outputs (called WR+XR+YR+ZR, these are the real-times-real portions) of
multipliers 2620, 2622, 2624 and 2626, and the
adder-subtractor-accumulator 2633 for the imaginary portion of the AZ
point in row-column array 2632 (called the AZI accumulator) gets loaded
with the sum/difference of outputs (called WI-XR-YI-ZR) of multipliers
2621, 2622, 2625, and 2626, respectively. By performing load operations at
clock C0 with no accumulation of the prior value (e.g., in AWR), the
previous values of the accumulators do not need to be zeroed. Note that,
since all four points for a single butterfly operation can be fetched
simultaneously from conversion RAM 2252, and the results of the respective
multiply operations must all be combined as they are formed, five-way
mixed add/subtract operations are provided for by each
adder-subtractor-accumulator 2633.
FIG. 84 shows the operations at each of the thirty-two
adder-subtractor-accumulators 2633 at a multiplier clock cycle command
denoted C1. For example, at C1, the adder-subtractor-accumulator 2633 for
the real portion of the AW point in row-column array 2632 (called the AWR
accumulator) gets loaded with the result of a five-way
subtraction/addition of the outputs (called AWR-(WI+XI+YI+ZI), these are
the imaginary-times-imaginary portions) of multipliers 2621, 2623, 2625
and 2627 and the prior contents of AWR The adder-subtractor-accumulator
2633 for the imaginary portion of the AZ point in row-column array 2632
(called the AZI accumulator) gets loaded with the sum/difference of
outputs (called AZI+(WR+XI-YR-ZI)) of multipliers 2620, 2623, 2624, and
2627, respectively.
Similarly, FIGS. 85 through 90 show the four-way and five-way operations
which take place at multiplier clocks C2 through C7, respectively. Table
2810 shows the order of calculation for one embodiment of a transposed
butterfly (calculating four radix-4 butterfly operations in eight
multiplier clock cycles), where each of the four points for one radix-4
butterfly are each in different sub-banks (e.g., one point in sub-bank
2290, one point in 2291, one point in 2292, and one point in 2293).
Twiddle-Factor Lookup Table 2610
In one embodiment, the twiddle-factor lookup table 2610 (also called a
sine-cosine ROM lookup table) comprises 512 fifteen-bit words, wherein
four words can be fetched in parallel. Each complex twiddle factor value
pair is fetched sequentially, wherein the first 15-bit word represents the
real part of the twiddle factor value pair and the second 15-bit word
represents the imaginary part, and four values are fetched simultaneously
(i.e., four real values, having 60 bits total, are fetched on an even
clock--e.g., clocks C0, C2, C4 or C6--and four imaginary values are
fetched on the following odd clock--e.g., clocks C1, C3, C5 or C7). In
another embodiment, the twiddle factor lookup table 2610 comprises of 256
thirty-bit words. The upper 15 bits represent the real part of the twiddle
factor whereas the lower 15 bits represent the imaginary part. Although
1024 complex-value pairs are required in order to produce a 1024-point FFT
or IFFT, the values are not unique, and the number of twiddle factors was
reduced by a factor of four by making use of the simple trigonometric
identities in mapping 360 degrees of twiddle factors to a 90-degree lookup
table. In one embodiment, the twiddle-factor lookup table was designed to
minimize DC offset caused by integer-based twiddle factors.
FIG. 92 shows a more-detailed block diagram of an
adder-subtractor-accumulator 2633. In one embodiment, multipliers 2620
through 2627 are each a 16-bit-by-16-bit multiplier. In one embodiment,
only the upper-order 16 bits of the resultant product are passed by MUX
2834. (In one embodiment, MUX 2834 is part of router logic 2634.)
Adder-subtractor 2833 performs a five-way addition/subtraction as defined
in FIGS. 83-90 and the two-way addition/subtraction as defined in FIGS.
75-82, under the control of sequencer 2640. In one embodiment, accumulator
2835 maintains enough bits above the binary point to accomodate overflow
bits and to provide an indication of overflow which does not get lost as
further addition/subtractions are performed on the accumulating data (in
other embodiments, one, two, or three such bits are maintained).
Input and Output Timing
Below is the detailed timing for one embodiment of the input and output
RAMs.
TABLE 7
Read Cycle for output RAM bank 2253
NOTE: The RAM clocks for the input and output banks are limited
to 10.24 MHz.
Symbol Parameters Condition Nom
Tcc Clock Cycle Time Pulse Minimum 20 ns
Width
Tchpw Minimum Positive CK Minimum 6 ns
Pulse Width
Tclpw Minimum Negative CK Minimum 6 ns
Pulse Width
Tavch Address valid to CK high Minimum 4 ns
Tchax CK high to address change Minimum 1.0 ns
Tchdox CK high to Data Output Minimum 2 ns
change
Tcd CK high to data valid Maximum 15 ns
Toe Output Enable time Minimum 0 ns
Toz Output Disable time Maximum 7 ns
TABLE 8
Write Cycle for input RAM bank 2251
NOTE: The RAM clocks for the input and output banks
are limited to 10.24 MHz.
Symbol Parameters Condition Nom
Tcc Clock Cycle time Minimum 20 ns
Tchpw Minimum positive CK Minimum 6 ns
pulse width
Tclpw Minimum negative CK Minimum 6 ns
pulse width
Tavch Address valid to CK low Minimum 4 ns
Tchax CK low to address change Minimum 1 ns
Twch .about.WE low to CK low Minimum 4 ns
Tchw CK low to .about.WE high Minimum 1 ns
Tdivch Data Input valid to CK low Minimum 4 ns
Tchdix CK low to Data input Minimum 1 ns
change
Tchdov CK low to Data Output Maximum 15 ns
valid
Tchdox CK low to Data Output Minimum 2 ns
change
Package for ASIC 2101
Package Dimensions and Pinout for one embodiment:
The ASIC 2101 generates 5 V TTL output levels and accepts 5 V CMOS or 5 V
TTL input levels.
TTL Input Levels are defined as follows:
VIL max=0.8 Volts
VIH min=2.0 Volts
CMOS Input Levels are defined as follows:
VIL max=0.2*VDD (0.9 V<VIL<1.1 V over 4.5 V to 5.5 V VDD range)
VIH min=0.7*VDD (3.15 V<VIH<3.85 V over 4.5 V to 5.5 V VDD range)
TTL Output Levels:
VOL max=0.4 Volts
VOH min=2.4 Volts
Functional Tests
Testing is broken down into a functional segment to verify device
functionality and a scan segment to test for faults in the physical
silicon. In one embodiment, the vectors are included in a test bench
compatible with LSI Logic's tools. The functions to be tested in the
device are listed below.
a. FFT/IFFT Operation--Vectors are provided which are characteristic of the
expected use in the system. The frequency domain vectors are passed
through an inverse transform (with appropriate bit scaling) and the
results stored. The vectors are then passed through a forward transform
(with appropriate scaling), and this final result analyzed These tests are
performed for the 1024-, 512-, and 256-point transforms. There are 10
frames of data for each test. The test bench includes 1024 vectors for the
1024-point transform, 5120 for the 512-point transform, and 2560 for the
256-point transform. There are twice this number of vectors passed through
the device to complete the test The total number of test vectors for this
test segment are about 36000.
b. FFT/IFFT Verification--A single sinusoid is passed through the 128-,
64-, and 32-point transforms. Both forward and reverse directions are
tested.
c. Bit Growth Tests. Each bit-growth pin (2117-2118) is exercised for the
1024- and 512-point transform in the forward and reverse direction.
d. Power Down Tests--The device is placed in the middle of a transform,
then powered down. The outputs are evaluated for correct state. The device
is then asked to perform a forward and reverse transform to validate that
the device can function after the reset.
e. Overflow Tests--An overflow condition is induced, and the device
evaluated for correct response (e.g. the overflow pin is actuated and the
event does not cause an adder to wrap around). The test includes an
overflow in the positive and negative direction.
f. Reset--The device is placed into the middle of a transform operation,
then reset. The outputs are evaluated for correct state. The device is
then asked to perform a forward and reverse transform to validate that the
device can function after the reset.
FIG. 93 is a high-level block diagram of one embodiment of modem receiver
2402 as shown in FIG. 71. The analog received signal-in is first processed
by bandpass-and-down-convert block 2750. In one embodiment, the analog
received signal-in is either 425 to 600 MHz or 550 to 770 MHz, and is
converted by bandpass-and-down-convert block 2750 to a signal which is 100
kHz on both sides of a 18.432 MHz center frequency. In one embodiment, the
tuning step size is 99 MHz. In one embodiment, analog-to-digital decimator
system (ADDS) 2850 (in one embodiment, this is a Sigma-Delta decimator
system that uses a Sigma-Delta analog-to-digital converter 2840) converts
this band-limited signal into decimated I and Q quadrature signals, each
15 bits wide, which have a symbol rate of 288 K symbols per second
(denoted 288 KBS), which are then processed by FFT block 2849. In one
embodiment, FFT block 2849 is equivalent to FFT system 2100 of FIG. 70.
The outputs of FFT block 2849 are then processed by post-processing block
2990 into digital data out.
FIG. 94 is a more detailed block diagram of modem receiver 2402. Analog
received signal in is fed to band-pass filter (BPF) 2740 which limits the
input signal to either 425 to 600 MHz or 550 to 770 MHz. The signal is
then amplified by variable-gain amplifier 2741, and mixed by mixer 2742
with a demodulator signal of either 627-802 MHz or 752-973 MHz generated
by signal generator 2747 as controlled by phase-locked-loop block 2746.
The mixer 2742 produces a difference-frequency signal centered at 202.752
MHz which is filtered by BPF 2743. The resultant signal is again
amplified, this time by variable-gain amplifier 2744, and then mixed by
mixer 2745 with a demodulator signal of either 221.184 MHz generated by
signal generator 2749 as controlled by phase-locked-loop block 2748. The
resultant signal is an analog signal centered at 18.432 MHz and having a
200 kHz bandwidth.
The resultant 18.432 MHz analog signal is then passed to an
analog-to-digital decimator which, in this embodiment, is denoted as the
components encircled by the dotted line referenced as Sigma-Delta
decimator system (SDDS) 2850'. The exact mix of components which are
included in SDDS 2850 can vary from embodiment to embodiment (i.e., where
the dotted line for SDDS 2850' is drawn may vary).
In FIG. 94, the 18.432 MHz analog signal is passed through BPF 2839 which
is centered at 18.432 MHz The 221.183 MHz signal is divided by divider
network 2838 to produce a 73.728 MHz signal which drives Sigma-Delta
converter 2840, and two 18.432 MHz signals (one of which is 90 degrees
shifted from the other) which drive digital I/Q detector 2841. Sigma-Delta
converter 2840 is any conventional Sigma-Delta converter, such as
described in An overview of Sigma-Delta Converters, by Pervez M. Aziz et
al., Vol 13, No. 1, IEEE Signal Processing Magazine, January 1996, which
is hereby incorporated by reference. Sigma Delta converter 2840 achieves
high resolution by oversampling the input signal at a frequency much above
the Nyquist frequency, and by providing a negative feedback path which
uses an analog representation, y.sub.a [n] of the quantized output signal
y[n]. FIG. 95 shows one such Sigma Delta converter 2840, having difference
block 2860 that forms u[n] which is input x[n] minus feedback y.sub.a [n].
Signal u[n] then is processed by discrete-time integrator 2863 and
quantizer 2864, which is simply a comparator. By providing a sampling
frequency which is high enough, the Sigma-Delta converter 2840 allows the
use of a 1-bit quantizer to achieve high overall resolution.
Referring back to FIG. 94, the 73.728 MHz quantized output of Sigma-Delta
converter 2840 is coupled to digital I/Q block 2841. In one embodiment,
digital I/O block 2841 is simply two 2-input AND-gates; one input of both
AND gates is connected to the 73.728 MHz quantized output of Sigma-Delta
converter 2840, the other input of the first and the second AND-gate is
coupled to the 0.degree. and the 90.degree. 18.432 MHz outputs of divider
2838. The outputs of digital I/Q detector 2841 are thus two serial
streams, that represent I and Q quadrature signals respectively. The
serial I and Q signals are then fed to coarse decimator and MUX 2842,
which converts the two serial streams into a single N-bit-wide
time-multiplexed I/Q parallel stream In one embodiment, this
time-multiplexed I/Q parallel stream is 10 bits wide. This
time-multiplexed I/Q parallel stream is clocked at 2.304 MHz, and provides
10 bits of I followed by 10 bits of Q, thus having 1.152 million samples
of I interleaved in time with 1.152 million samples of Q. This
time-multiplexed I/Q parallel steam is then fed to I/Q demux 2843 which
de-multiplexes the time-multiplexed I/Q parallel stream into separate I
and Q streams clocked at 1.152 MHz each, and each 10 bits wide. These
separate I and Q streams are then processed by three digital-processing
blocks: DC-offset adjust block 2844 that digitally adjusts for DC (direct
current) balance, DS_gain adjust block 2845 that digitally adjusts the
decimated signal gain, and DS_mix block 2846 that digitally adjusts the
phase.
The Sigma Delta decimator system (SDDS) 2850 is a N-bit A-to-D converter
which generates a one-bit serial data stream having resolution and
accuracy of N bits (in one embodiment, 15-bit resolution is obtained; in
another embodiment, the A/D has a 10-bit resolution with 9-bit linearity).
SDDS 2850 is running on the clock generator 2749 divided to 73.728 MHz
which oversamples the SDDS input signal in order that it only passes data
at the 18.432 MHz.+-.100 kHz, approximately. The following circuits
2841-2847 then take that 200 kHz of frequency that Sigma-Delta converter
2840 passes and shifts it down a base band, so basically it goes from 0 Hz
to 200 kHz. This a relatively slow signal and then SDDS 2850 turns that
into 10-bit parallel words. The Sigma-Delta converter 2840 is outputting a
serial 1 bit stream, which is ANDed with two 18.432 MHz square waves to
produce serial digital I and Q that are two 18.432 MHz gated square waves.
In one embodiment, the entire SDDS 2850 is integrated on a single VLSI chip
using 0.8 micron BiCMOS 4S+ technology fabricated by IBM Corporation, with
the analog circuits operating from a 5 volt supply voltage and the digital
circuits operating from a 3.3 volt supply voltage. This single-chip
implementation facilitates bit growth from stage-to-stage in order to
prevent or reduce truncation or round-off errors. Thus 10-bit I and Q
signals at the output of I/Q demux block 2843 are allowed, in one
embodiment, to grow to 25-bits at the output of DS_mix block 2846 through
the digital processing of DC-offset adjust block 2844, DS_gain adjust
block 2845, and DS_mix block 2846. For example, the N bits each of the I
and Q data streams at the output of I/Q demux block 2843 grow a little to
N.sup.+ bits after the digital processing of DC-offset adjust block 2844,
N++ bits after DS_gain adjust block 2845, and N.sup.+++ bits after DS_mix
block 2846. Decimators 2847 and 2847' select one out of every four values
from the I and Q data streams, respectively, thus producing a 288 kHz rate
of 25 bits each for the I and Q streams. These then pass through scaling
blocks 2848 and 2848' which scale each data stream to 15 bits, which are
denoted I' and Q' and are coupled to 15-bit FFT 2849.
One consideration with the large number of signals on such a single-chip
implementation is to minimize the number of different clock signals. In
the embodiments described for FIG. 94, for example, this is accomplished
by running a large number of blocks from a single clock, even though some
of those blocks might be able to run off a different and slower clock.
Overview of Data-Delivery Architecture
Referring now to FIGS. 96 and 97, there is shown the overall architecture
of a data delivery system 500 according to another embodiment of the
invention. The data delivery system 500 provides high speed data access to
the home 510 or office 512 (alternately referred to above in FIG. 1 as
remote units 46) over the HFC distribution network 11 using the
infrastructure of the modem-based telephony transport system 10 described
hereinabove. FIG. 97 illustrates the integration of the data delivery
elements in the HDT 12. The system allows users to have access to local
content 520 and the Internet 530 through services available at the cable
providers' premises or head end 32.
Among other things, system 500 provides: (1) user data access to the
Internet 530 and local content on a head-end server 520 through the
above-described access platform; (2) support for TCP/IP and transparent
bridging at the data link layer using a Dynamically Adaptive Transport
Switching (DATS) methodology (described below); (3) guaranteed, reliable
symmetrical data service at transfer rates from 64 Kbps to 512 Kbps, in
increments of 64 Kbps, for geographically dispersed individuals; (4)
guaranteed, reliable, symmetric shared access to a 8.192 Mbps data pipe
for geographically limited group of users connected in a routed
configuration; (5) mixing of data and telephony within a single HDT 12;
(6) network management for telephony, video, and data through integrated
CMISE and SNMP; (7) routed service through a head-end server, and (8) use
of HISU and MISU RF modem technology for transport.
Referring now to FIGS. 98-100, the Cable Data Modem (CDM) 535 for the
system 500 preferably can take three forms: a stand-alone box called the
Personal Cable Data Modem (PCDM) 540, a HISU add-in card called the Data
Modem Service Module (DMSM) 550, or a MISU add-in card called the Data
Modem Channel Unit (DMCU) 560. The stand-alone PCDM 540 has several
connector options; including standard F style coax cable, 10 baseT
Ethernet 542, and RS232544. The DMSM add-in card preferably resides in an
expansion slot on HISU 68 enclosure and will terminate a customer's
computer with a 10 BaseT Ethernet connection. The DMCU 560 add-in card
will reside in a line-card slot of the MISU and will provide a 10 BaseT
Ethernet routed connection to four users. Customers without an integrated
Ethernet port can add one to their system at minimal cost.
The system 500 provides connection to head end 32 services via the DATS
methodology. The DATS methodology intelligently allocates bandwidth in the
system to maximize both transport bandwidth and number of users while
providing guaranteed bandwidth. Under TR303/V5 switching environments a
central resource within the HDT 12 (described below) provides the
intelligence to allocate bandwidth and efficiently manage transport load.
This capability is implemented at the customer end by initiating a
connection to the head end 32 when data is available to send. When the
session is initiated, the head end 32 equipment determines the amount of
bandwidth to be allocated to the subscriber as configured during
pre-provisioning. The connection is maintained as needed and dropped when
transmission is complete. During the time that the connection is
maintained, bandwidth is guaranteed, providing the efficiency of packet
switching and the guaranteed bandwidth of connection switching. All
processing is performed through standard TR303/V5 call processing and
therefore integrates completely with telephony provided over system 500.
The architecture of the system 500 provides guaranteed bandwidth and
latency to all supported users. As indicated in FIG. 101, up to 24
concurrent users can be supported at 512 Kbps within a single 6 Mhz
transport channel.
Using the DATS methodology, bandwidth is dynamically allocated to support a
maximum of 240 users per 6 Mhz channel (e.g., see FIG. 13) at 64 Kbps.
Each user on the system is guaranteed the allocated bandwidth for the
duration of the session. As such, the bandwidth and latency seen by the
user remains constant, independent of the traffic load, application type,
or number of users. Assuming even distribution across all applications,
the average frame size carried downstream is 378 bytes, while the average
upstream frame size is 69 bytes.
The downstream latency from the HDT 12 is dominated by filling a 512 K data
pipe, all other latencies in the system are insignificant. Therefore on
average the latency to transfer a TCP/IP frame from the HDT to the
customer is 5.9 msecs. With Forward Error Correction (FEC) disabled, the
average latency in the upstream direction, from the customer end to the
HDT is also dominated by filling a 512 K data pipe. Therefore, on average
the latency to transfer a TCP/IP frame from the HDT to the customer is 1.0
msecs. Adding FEC will add up to 7 msecs to the upstream latency.
Data Transport and Framing
The system 500 provides transparent bridging and data transport. A
schematic representation of the data transport and framing is shown in
FIG. 102. Upstream data traffic begins at the customer end equipment 511
where frames are put out on the Ethernet link 542. The CDM 535 buffers the
frames and encapsulates them with HDLC framing 570. The HDLC frame 570 is
then sent out over the coax by a modem 101 as one or more of the possible
240 payloads on the coax. The frame, along with other data and telephony
streams is converted at the ODN 18 and sent to the head end 32 over
optical fiber. At the head end, the CXMU 56 demodulates and sends th e to
the CTSU 54, where it is routed to the appropriate LANU 580 as part of a
MARIO data steam. The LANU 580 buffers the complete frame, stripping the
HDLC framing prior to putting the frame on the local Ethernet. All routing
for the Internet is performed by the head-end server 590. The routing
services could also be incorporated in LANU 580.
In the case of downstream traffic, the head-end server 590 puts Ethernet
packets on the LAN. One to potentially seven LANUs 580 examine the
Ethernet address and selectively pick up packets destined for the
customer-end machines. Once the Ethernet packet has been accepted, the
packet is framed in HDLC and sent, via a MARIO stream, to the CTSU 54
according to the routing established when the call was initially
provisioned (explained further below). The CTSU 54 forwards the stream to
the CXMU 56 where it is sent over the HFC distribution network 11 to the
customer premises equipment 511. At the CDM 535, all packets received are
passed on to the local Ethernet connection 542. In the case of a broadcast
message, the LANU 580 at the head end will replicate the packet across all
attached connections. This allows the bridge to support protocols such as
ARP.
The system 500 will utilize the telephony error correction mechanism
described above with respect to system 10. Under the telephony error
correction scheme, forward error correction codes are generated for
upstream traffic but not for downstream traffic. Forward error correction
on upstream is generated at the ISU 100 (HISU 68 or MISU 66) and consumes
the 10th bit of each DS0, thereby protecting each DS0 separately. The
error correction can be disabled, but this is not recommended for data
transport The error detection/correction processing occurs on the CXMU 56
and data is delivered corrected to LANUs 580, DS1Us 48, in MARIO streams.
Therefore, the system 500 data architecture does not explicitly have to
deal with error correction The CRC of the HDLC frames provide for a level
of error detection above the error detection/correction of the CXMU 56.
Errors detected in the LANU 580 will be reported through the SNMP agent of
the LANU 580.
Component Design
The data delivery hardware for the head end 32 equipment consists of the
LANU 580 which interfaces with the HDT 12. The LANU 580 includes a board
responsible for all data transport to and from the head-end server LAN
591. In that function, the LANU 580 operates as a point-to-multipoint
connection that is responsible for concentrating up to 128 DS0s onto a
single Ethernet connection. The LANU 580 has the following features:
(1) It sits in a DS1U 48 slot and is form/fit compatible with the HDT 12
backplane.
(2) It provides 10 BaseT Ethernet connection to head-end LAN.
(3) It supports multiple LANUs 580 on a single LAN through a 10 BaseT
connection or fast Ethernet through hub.
(4) It concentrates up to 128 DS0 s into a single Ethernet Connection.
(5) It supports time slot assignment and super-channel aggregation across
four MARIO data steams independent of the CTSU 54.
(6) It supports Dual HDT 12 LAN ports.
(7) It maintains on-board FLASH for storage of operational code image.
(8) It supports Nx64 service for super-channels up to 512 Kbps.
(9) It guarantees time ordering of multi-channel calls independent of
transport system.
(10) It provides transparent bridging and broadcast of Ethernet frames
between head-end LAN 591 and CDM 535.
(11) It self-discovers Medium Access Control (MAC) addresses of CDM 535 and
filters Ethernet frames with on-board CAM.
A high-level block diagram of the LANU 580 is shown in FIG. 103. The LANU
580 preferably employs a single processor, unified memory design In this
embodiment, the main processor on the LANU 580 is the Motorola MC68MH360
QUICC32 (581), running at 33 Mhz, which can deliver approximately 5 MIPS.
The QUICC32 is actually a dual processor embedded controller that also
contains a dedicated independent RISC processor communications called the
CPM. The CPM along with Direct Memory Access (DMA) can access memory and
move communication traffic without placing any performance burden on the
main processor. In addition to acting as the host processor, the QUICC32
can also act as a slave. The LANU 580 implementation is preferably based
on 4 QUICC32s with one acting as master and the remaining three operating
in slave mode. The combination of the four QUICC32s provide the majority
of the system peripherals.
There are several types of memory 582 on the LANU 580. The first and
largest is 8 MBytes of page-mode DRAM for the storage of LANU operational
code. In addition to operational code, the DRAM also holds the routing
tables, data buffers, and buffer descriptors needed to process data
traffic. Second, the LANU contains 2 MBytes of FLASH memory to store an
image of its operational code. On power-up, the image is loaded into DRAM.
Finally, the LANU also contains 128 Kbytes of EPROM to hold boot code. The
boot code will load the contents of the FLASH into DRAM on power-up. If
greater performance is desired, fast SRAM can be added to hold the data
buffers and buffer descriptors for the data traffic.
There are two sets of system peripherals on the LANU: those that are
contained within the QUICC32s and those that are not. Regardless of
location, peripheral control on the LANU 580 is managed by the System
Integration Module (software) (SIM) of the master QUICC32. The SIM is
responsible for managing generation of Chip Select (CS) and other control
lines.
The most important of the system peripherals are the QMC HDLC controllers
586 located in the QUICC32s. The QMC of the QUICC32 can implement up to 32
HDLC controllers running at 64 Kbps and are used to process the data
traffic in the MARIO streams from the CTSU 54. The QMCs 586 can also
support super channels by aggregating multiple 64 Kbit channels into
higher data rate channels. The QMC 586 is implemented in the CPM of each
QUICC32 and with the addition of SDMA can autonomously move communications
data to and from main memory 582. Operation of the QMC 586 is controlled
by the master processor 581 through the use of buffer descriptors that
reside in external memory 582.
In addition to the QMC 586, three additional HDLC controllers. 587 are used
to provide two 2 Mbps HDT backplane LAN ports and a 10 Mbit Ethernet port.
As with the QMC 586, the master processor 581 controls the HDLCs through
the use of buffer descriptors, but unlike the QMC, these data structures
reside in dual ported RAM (DPR) that are internal to the QUICC32s.
There are two additional system peripherals that reside outside of the
master and salve QUICC32s. The first is the Rate Adaptation/Ninth Bit
Signaling/Time Slot Assigner logic block (RA/NBS/TSA) 588. The RA/NBS/TSA
588, as its name suggests has three functions. The rate adaptation
function is responsible for converting the 2.56 Mbps, 9 bit data format of
the input MARIOs to the 2.048 Mbps, 8 bit data format of the QMC 586 in
both the receive and transmit directions. In both directions the RA is
also responsible for managing any information that may be placed in the
9th bit.
The NBS is responsible for transmitting and receiving the data that is
carried with each DS0 in the ninth bit The ninth bit is used for two
distinct functions. In the upstream direction the ninth bit carries
information regarding the ordering of data within a multi-channel call.
The signaling consists of a repeating number that indicates which time
position the DS0 occupies in the multi-channel call. The format of the
upstream ninth bit signaling is shown in Table 9.
TABLE 9
Upstream Ninth Bit Signaling
Bit Contents Description
1 "1" Sync Pattern
2 "1" Sync Pattern
3 "1" Sync Pattern
4 "1" Sync Pattern
5 "0" Sync Pattern
6 "0" Sync Pattern
7 D[8] Order Number, Bit 8 [MSB]
8 D[8]* Inv. Order Number, Bit 8 [MSB]
9 D[7] Order Number, Bit 7
10 D[7]* Inv. Order Number, Bit 7
11 D[6] Order Number, Bit 6
12 D[6]* Inv. Order Number, Bit 6
13 D[5] Order Number, Bit 5
14 D[5]* Inv. Order Number, Bit 5
15 D[4] Order Number, Bit 4
16 D[4]* Inv. Order Number, Bit 4
17 D[3] Order Number, Bit 3
18 D[3]* Inv. Order Number, Bit 3
19 D[2] Order Number, Bit 2
20 D[2]* Inv. Order Number, Bit 2
21 D[1] Order Number, Bit 1
22 D[1]* Inv.Order Number, Bit 1
23 "0" Sync Pattern
24 "0" Sync Pattern
The numbers can range from 1 to 128 with a 0 indicating that the DS0 has
not been assigned a position. During a call, the processor 581 will
monitor the ninth bit signals from all the channels and once the order has
been established, the processor will configure the TSA function to order
the channels. Once the order has been established, the processor will
periodically monitor the ninth bit signaling to detect any changes in
ordering (i.e., frequency hoping due to excessive errors).
In the downstream direction, the NBS is used to enable data transmission.
Once the LANU 580 receives the ordering information for the channels, the
processor will enable transmission by sending a data pattern over the
downstream ninth bit of the first DS0 of a multi-channel call. The format
for the "Data Dial Tone" signaling is shown in Table 10.
TABLE 10
"Data Dial Tone" Downstream Ninth Bit Signaling
Bit Contents Description
1 "1" Sync Pattern
2 "1" Sync Pattern
3 "1" Sync Pattern
4 "1" Sync Pattern
5 "0" Sync Pattern
6 "0" Sync Pattern
7 CMD[15] Command Bit 15, MSB
8 CMD[14] Command Bit 14
9 CMD[13] Command Bit 13
10 CMD[12] Command Bit 12
11 CMD[11] Command Bit 11
12 CMD[10] Command Bit 10
13 CMD[9] Command Bit 9
14 CMD[8] Command Bit 8
15 CMD[7] Command Bit 7
16 CMD[6] Command Bit 6
17 CMD[5] Command Bit 5
18 CMD[4] Command Bit 4
19 CMD[3] Command Bit 3
20 CMD[2] Command Bit 2
21 CMD[1] Command Bit 1
22 CMD[0] Command Bit 0, LSB
23 "0" Sync Pattern
24 "0" Sync Pattern
Two commands are defined. The first is "Idle Sync" (bit pattern for
CMD[15:0] is 0000 0000 0000 0000) is sent during idle times to synchronize
the receivers. All idle syncs sent from the LANU 580 will be synchronized
themselves to simplify the reception of order information in the upstream
ninth bit signaling. The second command is "Data Dial Tone" (bit pattern
for CMD[15:0] is 1000 0000 0000 0000) and is sent once the order of the
super-channel has been established. This command will instruct the CDMs to
enable transmission.
The TSA is responsible for ordering and aggregating channels that are sent
to the QMCs 586. Although TSA has the ability to aggregate up to 128
channels into a single data stream, most applications will aggregate
multiple super-channels, up to 8 DS0s per super-channel, among four
DS1-like channels feeding the four onboard HDLC controllers. For 1 to 32
DS0 data streams, the DS0s are aggregated and sent to the QMC 586 as a
2.048 Mbps serial stream. Within a single DS1-like stream, multiple
super-channels can be supported by connecting to a single QUICC32 QMC 586.
The TSA can also aggregate 128 DS0s into a single 8.192 Mbps data stream
and connect it to a QUICC32 HDLC controller. In both cases the TSA is
configured to insure that the time ordering of the data is maintained.
The final peripheral that does not reside in a QUICC32 is the Contents
Addressable Memory (CAM) 589. The CAM performs memory accesses based upon
data rather than address and is used to quickly determine whether an
Ethernet frame should be accepted. The Ethernet controller 587a interfaces
to the CAM 589 through glue logic and the reject input. When a frame is
received that is not in the CAM, the CAM logic asserts the reject control
line and the received portion of the frame is discarded. The buffer depth
of the Ethernet controller is set so that no memory accesses are generated
on rejected frames. The CAM is available off the shelf from MUSIC
semiconductor.
The LANU 580 sits in a DS1U 48 slot at the HDT 12 and is form and fit
compatible with the DS1U 48 to be backplane compatible. The LANU 580 has
four major system connections: MARIO 592, Ethernet 593, HDT LAN 591 and
clocking (input not shown in FIG. 103). The four MARIO connections connect
the LANU to the CTSU over the backplane. The four MARIO connections
provide up to 128 bidirectional, 64 Kbit channels. The Ethernet connection
connects the LANU to a local 10 Mbit, 10 BaseT LAN. The 10 BaseT
connection will take the place of a single T1 connection of a DS1U. The
connection is terminated by wire-wrapping the 10 BaseT to the backplane
and routing to a patch panel. For call provisioning and other system
functions, the LANU 580 connects to two common 2 Mbps HDLC LANs on the
backplane. Finally, system-clocking is provided by an external source such
as a Building Integrated Timing Supply (BITS).
Functional Description of Upstream Communications
For upstream traffic, the HDLC framed data available on the MARIO interface
passes through the rate adaptation and TSA block 488. In this block, the
2.56 Mbps MARIO interface is rate adapted down to 2.048 Mbps. As part of
rate adaption, the ninth bit of each DS0+ of the MARIO stream is stripped
and sent to the NBS logic. The ninth bit carried an order number that is
used to time order the DS0s in multi-channel calls. Once the order numbers
are established, the processor 581 configures the TSA to re-order the
multi-channel calls and target the super-channel to a QMC 586. For
super-channels composed of 32 or less DS0s, the call is placed in a single
2.048 Mbps data stream, along with other calls and sent to a QMC 586. For
128 DS0 calls, the DS0s are placed in a single 8.192 Mbps stream that is
target to a QUICC32 HDLC controller 586 configured for 8.192 Mbps HDLC.
Whether targeted to a 2.048 Mbps QMC or 8.192 Mbps HDLC, the frames are
accumulated and transmitted on the local Ethernet LAN.
Functional Description of Downstream Communications
For downstream traffic, data on the LAN is filtered according to the
destination MAC address. If the MAC address is in the Content Addressable
Memory (CAM) 589, the LANU 580 will accept the Ethernet frame. Once the
frame is accepted, the LANU 588 accesses a routing table in memory 582 to
select the appropriate MARIO slot for transport The frame is then
scheduled for transmission and the HDLC controller 586 takes over. In the
downstream direction for 32 DS0 or less calls, the HDLC controller 586 is
responsible for creating the MARIO stream and encoding the data into HDLC
format. For 128 DS0 calls, the 8.192 Mbps HDLC data stream is split among
the four MARIO interfaces (A-D). The time ordering and aggregation in the
downstream direction is controlled by the TSA. After the data passes
through the TSA, the ninth bit signaling information is added to indicate
that data transmission is enabled. At the same time that the ninth bit
signal is added, the data stream is rate adapted up to the 2.56 Mbps rate
of the MARIOs.
For data movement the on-board memory 582 provides the switching and
buffering necessary to move traffic from the Ethernet to the MARIO and
vice-versa. The data switching provides a virtual circuit based upon
routing information constructed during call provisioning that maps the MAC
address to MARIO time slot. The LANU 580 constructs this routing table
from information provided during call provisioning and by examining the
source MAC address of Ethernet frames received from the customer end. The
MAC addresses supported by the LANU are then placed in the CAM 589 where
they can be used to filter Ethernet traffic.
In addition to the MARIO and Ethernet interfaces, the LANU 580 also
supports two HDT backplane LAN 591 interfaces at 2.048 Mbps. The HDT LAN
is maintained by the SCNU 58 at the head end 32 and is used to pass
configuration information to the LANU 580. The combination of LANU 580,
call processing, and pre-provisioning will provide the MARIO route (time
slot in the MARIO data stream) to all customer devices.
Customer Premises Equipment (CPE)
The CPE of the system 500 comes in different implementations. Although the
basic data framing and transport remain the same across all the
implementations the underlying modem technology and form factor differs.
There are currently three general types of CPE defined: HSU based, MISU
based, and Stand-alone (a variant of the HISU based implementation). Each
of these implementations are discussed separately below.
Data Modem Service Module
A simplified block diagram of the DMSM is shown in FIG. 104. DMSM 550
supports up to 8 DS0's for data traffic. Data is interfaced to the HISU 68
through a Service Growth Module (SGM) interface that implements the bridge
router functions of the data connection. The bridge/router is based upon a
68302 embedded processor 605 with 2 MBytes of DRAM 604 and 256 Kbytes of
FLASH memory 606. One embodiment of the customer interface is a 10 BaseT,
10 Mbit Ethernet connection 602. The interface between the bridge/router
and the HISU modem is a Super-LUIGI (SLUIGI) interface operating at 2.048
Mbps. Data in the upstream direction is accepted by the Ethernet
connection and relayed to the HISU 101 modem over the SLUIGI interface as
HDLC encoded data. In the downstream direction, data from the SLUIGI
interface is passed to the HDLC controller 607 and then sent out the
Ethernet connect through Ethernet controller 609. System memory consists
of 2 Mbytes of DRAM 604 which contains the operational code that
implements the bridge/router functions as well as an SNMP agent The 256
KBytes of FLASH 606 is used for storage of the operational image and can
be updated with TFTP transfers.
In the upstream direction, the HISU interface logic 608 is responsible for
generating the ninth bit signaling information for each DS0 and
interfacing with the SLUIGI stream of the HISU. The 2.048 Mbps serial data
from the HDLC controller is stuffed in the appropriate time slots of the
SLUIGI interface. In the case of 64 Kbit traffic, all HDLC data is placed
in a single SLUIGI time slot In the case of multi-channels calls, all HDLC
is placed in adjacent SLUIGI time slot receiving the first byte of HDLC
data In addition, the interface logic will generate an order number for
each DS0 with "1" assigned to the first DS0 and "8" assigned to the last
DS0, "0" is reserved to indicate that the position of the DS0 has not been
set In the event of frequency hopping, the interface logic will continue
to number the DS0 time-slots in the order that they are received. For
example, under normal circumstances DS0 ninth bit signaling would tag the
time slots as "12345678-" for an eight channel call. If DS0 4 is lost due
to ingress, then the new order would be "123-45678." This signaling
information is used at the head-end to reorder the DS0s independent of the
frequency hopping and transport services.
In the downstream direction the HISU interface logic 608 is responsible for
taking data from the SLUIGI interface from the RF modem 101 and giving it
to the HDLC controller. In addition, the HISU interface logic monitors the
ninth bit signaling information of the first DS0 to detect the "Data Dial.
Tone" sequence. The "Data Dial Tone" sequence is sent by the head-end to
enable data transmission. During the call provisioning process the HISU
interface logic sends the ninth bit ordering information as soon as the
SLUIGI time slots indicate that they can accept data. It is not until the
HISU interface logic gets a positive acknowledgment through the "Data Dial
Tone" that data is sent to the head-end in the upstream direction.
Data Modem Channel Unit (DMCU)
A simplified block diagram of a DMCU 610 is shown in FIG. 105. The DMCU
supports up to 128 DS0s for data traffic. The data interface to the MISU
is a specialized channel that sits on the MISU backplane.
The basic design of the DMCU 560 is very similar to the design of DMSM 550.
The most notable difference is the interface to the RF portion is two
8.192 Mhz serial channels. This allows the MISU interface to support a
symmetrical 8.192 Mbps Ethernet connection. Because of the higher
throughput the MISU is based on the MC68360 614 that can support both the
8.192 Mbps HDLC connection as well as the 10 Mbit Ethernet In front of the
10 Mbit Ethernet interface 612 is a router 616 that allows four users
access to the 8.192 Mbit data connection. The router 616 design insures
security for all connected users. The design contains 2 Mbytes of DRAM 618
and 256 bytes of FLASH 619. Like the HISU design, the FLASH can be
remotely updated with TFTP.
The DMCU 610 has an equivalent interface function that moves data from the
HDLC controller 611 to the RF modem and works in a very similar way to the
DMSM 600 with the exception that the MISU modem interface is formatted as
two SLUIGI streams that are clocked at 8.192 Mbps. Between the two SLUIGI
streams; 128 DS0s can be carried between the HDLC controller 611 and the
modem. The MISU interface logic 613 is responsible for buffering and then
sending data over the dual S-LUIGI interface to and from the RF' modem. In
the upstream direction the MISU interface logic 613 generates order
numbers for each of the 128 DS0s over the ninth bit of the DS0+. The order
numbers generated on the MISU work in the same way as they do on the HISU.
In the downstream direction, the MISU interface logic 613 is responsible
for moving data from the S-LUIGI streams to the HDLC controller 611. The
MISU interface logic also monitors the ninth bit data stream from the
first DS0 to detect the "Data Dial Tone" that enables data transmission.
Ethernet controller 617 moves the data to the router 616.
Stand-alone Data Modems
The stand-alone data CDMs are based upon the HISU design In the stand-alone
designs, the RF modem 101 of the HISU is tightly integrated with the
bridge/router design Like the DMSM 600, the stand-alone supports from 64 K
to 512 K. The interfaces are identical with several options: standard "F"
style connector to the cable, 10 Mbit Ethernet and RS232 connection to the
customers equipment.
System Software and Call Processing
In the TR-008/V2 system, calls are provisioned and nailed up at time of
installation. Under this scenario an operator at the head-end 32 is
responsible for determining the MARIO configuration and transfer rates (64
K to 512 K). The DATS methodology of present invention utilizes TR-303/V5
call processing to provide dynamic allocation of bandwidth. To maintain
the telephony oriented architecture of the access platform of the present
invention, the LANU 580 takes on responsibility of a limited subset of the
Central Office (CO) functions. This approach has the distinct advantage
that the data sessions are fully integrated with telephony.
At the time of deployment, a LANU 580 will be identified as a "Master" LANU
(mLANU). The mLANU will have the responsibility to maintain the CO-like
functionality for all data calls to the HDT 12. In order to perform these
functions, the mLANU will represent itself to the HDT as an IDT. When the
mLANU is pre-provisioned, the mLANU will be given an IDT identifier and
assigned a Time Slot Management Channel (TMC) channel from the CTSU 54.
Regardless of the number of LANUs 588 in the HDT 12 a single MLANU will
allocate and keep track of available DS0s for all the LANUs in the HDT. As
customers are configured, a Call Reference Value (CRV) for the selected
CDM will be assigned to identify the customer. The CRV along with the
number of data channels will be added to a call provisioning database on
the MLANU.
The call processing sequence for call origination is shown in FIGS.
106-109. Call processing begins when the CDM generates an "Off-Hook"
message over the IOC associated with the HISU, MISU or Stand alone CDM
(described above). After the "Off-Hook" message is received at the CXMU 56
then sends a "Request Service" message over the backplane. LAN 591 to the
CTSU 54 identifying the CRV of the originator. After receiving the
"Request Service" message, the CTSU 54 sends a set-up message to the mLANU
580 over the TMC (DS0 in a MARIO stream). The MLANU uses the CRV to access
the on-board database in memory 582 and determine the number of DS0s to
allocate the call. Once the number of channels has been determined, the
mLANU identifies the DS0 and DS1 for the call. The mLANU then sends a
"Make Cross Connect" message to the CTSU 54 over the TMC identifying the
DS0 and DS1 and their association with the CRV. In response the CTSU 54
sends a "Req. Bandwidth" message to the CXMU 56 over the backplane LAN 591
to allocate the bandwidth in the transport
Preferably, the DAT methodology and system provides that each subscriber is
represented in the database as having subscribed to a certain level of
bandwidth per data connection. For example, a subscriber may sign up for
512 K of bandwidth. Upon call setup or connection, the DAT methodology
thus assigns each user the number of channels required to achieve the
subscriber's bandwidth. However, in certain cases, the transport system
500 may not have the necessary bandwidth to allocate to a subscriber their
normal subscripted bandwidth. Under these circumstances, the subscriber is
allocated a lesser amount of bandwidth, for example 64 K of bandwidth. By
dynamically adjusting the amount of bandwidth assigned at call setup for
each call, the MLANU can maintain a minimum level of bandwidth for each
subscriber. However, existing constraints prevent small decremental
bandwidth reallocations. Constraints existing within the system 500 as
described above will enforce the halving of bandwidth of some subscribers
to accommodate additional subscribers, rather than by means of a more
evenly distributed loading. As shown in FIG. 107, as the number of
subscribers increase (on the horizontal axis) any given user's allocated
bandwidth will halve at a certain user density. The spread around the
average indicates that some users will necessarily lose half their
existing bandwidth earlier under loading than will others. The
constraining factor in preventing a more equitable burden is the window
nature of the present HISU 68 RF tuning. As noted above, the HISU tunes to
one of 24 IOC channels spread throughout the 6 MHz cable channel and has
access to five payload channels above and five payload channels below the
selected IOC frequency. It cannot borrow payload channels outside this
window of ten channels, so therefore there is no way for the "25th" user
to borrow just one channel each from seven other users. It can only sit on
top of one of the existing IOC payload windows and takeover half of the
window bandwidth. At that point, 23 users would be granted 512 kbs
bandwidth and two users would each get 256 kbs bandwidth The "26th" user
would result in 22 users with 512 kbs and four users with 256 kbs, and so
on. This general pattern is repeated at a load of 72 users and 120 users.
(There is a discontinuity in the pattern from 48 to 72 users due to the
previously unused two of ten DS0's per window being pressed into two DS0
128 kbps service.) The graph of FIG. 109 illustrates the distribution of
bandwidth to users as the number of users increases. It is also
contemplated that a subscriber could have different default or standard
data rates depending on the time of day or day of week, or based on system
loading, such that a user can receive even more bandwidth than their
standard rate under certain system loading conditions, such as if the
system is loaded below a predetermined threshold at the time the
subscriber seeks a connection.
Also, using the ninth bit signaling, the MLANU can "steal" bandwidth from
other (e.g. high capacity) users. This is done by removing the Data Dial
Tone from a subscriber using the ninth bit signaling. This quiesces the
user's line, allowing, the number of channels for that user can be
reassigned to increase the bandwidth allocated to the user. This technique
is also be used to decrease the number of channels assigned to a user.
In order to establish the transport, the CXMU 56 trains the modems (as
described above with respect to system 10) and associates the available
tones with DS0s. Once the training is complete, the CXMU 56 sends a "Pass"
message to the CTSU 54, which in turn informs the mLANU over the TMC that
the call is complete with the "Call Complete" message. In response, the
MLANU 580 configures the HDLC controllers 586 and the TSA 588 on the mLANU
or another LANU through communications over the backplane LAN. At this
point the pipe is established but data is not yet enabled.
In order to actually begin data transmission two additional steps have to
occur across the ninth bit signaling of the DS0s. At the point where the
modems are trained the HISU 68 or MlSU 66 interface logic (608, 613) will
be enabled to transit data Once the transmit is enabled, the interface
logic will begin transmitting the DS0 ordering number in the ninth bit of
each DS0. At the LANU 580, the processor 581 will monitor the ninth bit
signaling to determine when all DS0s have established their order. Once
all DS0s have established order, the LANU 580 will send the "Data Dial
Tone" pattern on the ninth bit of the first DS0 in the multi-channel call.
When the ISU 100 receives the "Data Dial Tone" data communications are
enabled and data transmission begins.
A session is terminated at the customer end when no data is available for
transmission by generating an "On Hook" message. The call processing
sequence for an "On Hook" message is shown in FIG. 110. When the CDM
terminates the connection, an "On Hook" message is sent over the IOC to
the CXMU 56. The CXMU 56 is response sends an "On Hook" message,
identifying the CRV, to the CTSU 54. The CTSU 54 then sends a "Tear Down"
message to the mLANU 584 over the TMC. At the MLANU 580, the connection is
deleted from the connection database and then released. If the connection
is not on the mLANU, the mLANU will send a "Release Channel" message to
the target LANU 580 and also will send a "Release Cross Connect" message
to the CTSU 54. The CTSU 54 will release the cross connects used for the
connection and then send a "Release Bandwidth" message to the CXMU 56. At
the CXMU 56 the mapping between tones and DS0s is lost and the connection
is lost. When the connection is lost, the CDM will lose the "Data Dial
Tone" in the ninth bit signaling of the first DS0 of the call.
The LANU 580 can also be configured to bring up connections to customer-end
equipment This allows for notification of incoming e-Mail and personal Web
pages without tying up idle bandwidth To do this, the master LANU in each
system will maintain a mapping between the MAC address for each data
element in the system and cross that with the CRV. Then an Ethernet packet
is put on the head-end LAN 591, and a LANU 580 will read its MAC address
and determine whether the connection is up to the device. If the
connection is up, the packet will be forwarded over the HFC transport. If
the connection is not in place, the receiving LANU 580 will generate a
connection request to the mLANU. The mLANU will then signal the transport
system over the TMC to bring up a connection to that device using the IOC.
The receiving LANU 580 will then send the data once the connection has
been established.
LANU 580 Software
The LANU software 620 is responsible for the all major function of the data
concentration of the head-end equipment. A simplified schematic diagram of
the LANU software is shown in FIG. 111. The software 620 of the LANU
consists of three major components: bridging 621, HDLC LAN manager 622,
and data IDT 623. All three tasks will operate as applications on top of
the embedded controller operating system "pSOS" kernel in the processor
581. The pSOS kernel will provide the base for the multi-tasking operation
of the software 620.
The most important task to the actual transport of data will be the
bridging task. The bridging task has several functions. First, the task
will be responsible for providing the virtual switch between the MARIO and
Ethernet interfaces. The task will be implemented as in "interrupt on
receive" task that will execute at interrupt level. At either interface,
an interrupt is issued when an entire frame has been received and stored
in buffer memory (FIG. 103, 582). During the interrupt service routine,
the packet will be handed off by modifying the associated buffer
descriptor after looking up the routing in the bridging table (stored in
memory 582). For upstream traffic (HDLC to Ethernet), the source of the
first packet will be read to discover its MAC address. This address will
be added to the bridging table and written to the Ethernet CAM 589 for
filtering.
A second function of the bridging task is the creation and maintenance of
the bridging table. The bridging table will match the MAC address of the
CDM 535 with the MARIO DS0s so that data can be moved between the Ethernet
and HDLC. During call processing, the DS0s that are allocated to the call
will be identified by the mLANU 580 through backplane LAN (591) Messaging
and installed in the bridging table. As described above, when the first
frames begin to flow from the CDM, the source MAC address will be
identified and the table entry for the CDM will be complete. At this point
data will flow in both directions. Once the MAC address has been
discovered, the bridging table entry will remain intact until the
connection is terminated by the CDM.
A third function of the bridging task is maintenance of an SNMP agent. SNMP
traffic will be handled and processed from the Ethernet port. The agent
will support a standard MB-II information database for transparent
bridging. In addition, objects will be added to the MIB that are specific
to the data architecture to facilitate CMISE-SNMP integration and
different billing options.
Finally, the bridging function may support link-layer encryption/decryption
on the bridged data. Encryption/Decryption may be software only or
hardware assisted depending upon the desired performance of the system. In
either case, this function will execute as an application on top of pSOS.
Another component of the LANU software 620 is the HDT LAN manager 622. The
HDT 12 LAN 591 is used to communicate system messages between the elements
of the HDT 12. During pre-provisioning, the SCNU 58 will communicate
system parameters such as CRV, IDT ID, and number of channels accessible
by the CDM to the mLANU 580. These parameters will be used in the
construction of the call provisioning table resident on the mLANU. During
call provisioning, the mLANU will examine the provisioning table for
available DS0s and use the HDT LAN 591 to set up MARIO configurations on
other LANUs 580.
Another important function of the HDT LAN manager software 622 is support
for field software upgrades. During download, the LANU 580 will take the
image from the HDT LAN and store it in on-board FLASH memory. Aside from
the SCNU 58, the LANU will be the only board in the HDT 12 that will load
its image from its own FLASH on power up. As such, support for image
download from the SCNU 58 Ethernet port will need to be added to the SCNU
software. A final function of the LAN manager software 622 is to provide
the network management access to the SNMP environment of the LANU.
The final major task of the LANU software 620 is the data IDT 623. As
described above, the system 10 of the present invention is designed to
provide access from POTS to a CO switch. As such, the burden of resource
allocation and assignment of DS0 terminates at the switch. Since the data
architecture of the system 10 terminates at the head-end there is no such
centralized resource in the architecture to provide the services of the
switch. In order to provide the services required to terminate the data
"calls" a single LANU 580 functions as the data IDT.
The function of the data IDT is to provide a single point of reference for
the data resources of the HDT 12. During pre-provisioning of the LANU
hardware a LANU 580 will be designated the "Master" LANU (mLANU) and
assigned an IDT identifier. The mLANU 580 will then take on the function
of the switch for data calls by maintaining a table that maps CRVs to
service level (# of channels). In addition, the mLANU will maintain a map
of all available DS0s an all LANUs (including the mLANU itself) installed
at the HDT. A copy of the call provisioning table will be kept in on-board
FLASH so it can survive a power loss.
In order to maintain compatibility with standard telephony traffic, the
means of communication between the CTSU 54 and mLANU will be a TMC
connection over one of the DS0s within a MARIO. During call provisioning,
the CTSU 54 sends a setup message over the TMC and the mLANU will respond
with a "Make Cross Connect" message that identifies the DS0, DS1, and CRV
for the connection. As discussed previously, the mLANU will also configure
the LANU 580 for the connection through communications over the HDT 12 LAN
591. Therefore the data IDT software 623 will emulate the switch though
its communications with the CTSU 54 over the TMC using Q.931 compatible
messaging.
On all LANUs in the HDT 12, whether master or not, the data IDT software
623 will be responsible for configuring the TSA and communicating with the
bridging task. In configuring the TSA, the data IDT will monitor the
sequence numbers in the ninth bit signaling and appropriately configure
the interconnect so that the order within a multichannel call is
maintained. In addition, the data IDT software 623 will communicate the
state of the connection to the bridging task to open up the data pipe.
Another important function of the data IDT software 623 is to provide the
information needed to provide billing and other accounting functions. As
an IDT like function the data IDT will also support standard CMISE
objects.
CDM Software
Software provided in a CDM 535 (executing on the local processor 605 or 614
and represented by such elements) will provide the same types of
functions. The major function of the CDM software is to provide the
bridge/router (brouter) functionality at the customer-end. In supporting
the brouter function the CDM software supports IP routing, PPP, and SLIP.
As part of IP support the CDM supports TFTP for downloading new code
images. The CDM also supports a standard SNMP agent with a full MIB-II
information base. Preferably, the software executes on either the 68302
(605) or 68360 (614) processor.
The control of the modem portion is with the standard MISU or HISU software
running on a Motorola 68HC11. This code supports all the alarm conditions
and communications set out above for IOC communications. The interface
between the RF modem and the brouter is preferably a hardware only
implementation.
Network Management
Network management of the data architecture of system 500 is preferably
provided by both CMISE and SNMP. The CMISE portion of network management
will be responsible for the transport mechanism for data, while SNMP will
be used for data network oriented management. In this environment, SNMP is
an overlay to the CMISE environment.
As with all telephony services, the data architecture will depend upon
CMISE for network management of call provisioning and other transport
related functions. In addition, CMISE will be responsible for accounting
on data connections. This approach provides for a very flexible billing
system where services can be billed per connection time, bytes passed, or
packets passed. Statistics will be collected in the mLANU and reported to
the network manager.
SNMP management is used to provide data services management for the data
architecture. In this way the data architecture will resemble a standard
data network. Within SNMP management, the LANU 580 and CDM will maintain
SNMP agents compliant with the MIB-II standard. In order to support an
SNMP agent both the LANU and the CDM will need to support the UDP and IP
protocols in addition to the SNMP protocol. In order to provide a single
point of management for data and telephony, both CMISE and SNMP are
preferably integrated into the same element manager. This level of
integration will simplify billing by providing several options such as
bill by connection time, bytes passed and packet passed.
Asymmetrical Data Delivery
For many casual residential users, data traffic can be characterized as
mostly "bursty" (intermittent), downstream traffic with relatively, small
upstream needs. The most cost effective means of delivering services such
as Web browsing, file downloads, and CD-ROM preview is asymmetrical
transport. The asymmetrical data transport embodiment of the invention
includes a customer premise unit or Personal Cable Data Modem that
contains a 30 Mbps, QAM downstream demodulator (PCDM-30) 620a, as shown in
FIG. 112. PCDM-30 also includes an OFDM upstream modulator supporting a
minimum of 64 Kbps guaranteed, non-shared bandwidth. The connection to the
customer-end equipment is 10 BaseT Ethernet that supports standard TCP/IP.
At the head end 32, an ASMU 622a supports multiple users on a single 30
Mbps channel which occupies 6 MHz of spectrum outside of the channels of
the telephony transport system 10. In addition to the downstream
modulator, the ASMU 622a concentrates the return channels by interfacing
with the HDT 12. Upstream traffic is carried over as a single DS0 and
integrated with the downstream transport on the ASMU 622a. The connection
from the ASMU to the head-end services is 10 BaseT, but higher capacity
industry standard connections are also possible.
The ASMU 622a sits at the head end 32, but not in the HDT 12. The function
of the ASMU 622a is to integrate the upstream path for up to 400 DS0s
(Configurable from 64 Kbps to 512 Kbps) and a 30 Mbps shared downstream
Each LANU 580 will generate an 8.2 Mbps HDLC stream that contains the 64
Kbps Ethernet packets from all the users that are attached to LANU through
the transport system. On the ASMU 622a, up to four of these, are
aggregated, and sent out to the head services over 10 BaseT Ethernet. In
the downstream direction, the data on the 100 BaseT is filtered on the
ASMU, and those packets destined for the customer end products are
accepted and then modulated onto the 30 Mbps shared medium.
In order to register a modem, the customer-end modem sends out an IP packet
to identify itself. This causes the LANU 580 to assign an HDLC address
that is mapped to the MAC address of device. This information is passed to
the ASMU 622a so that the HDLC address can be used by the modulator over
the HDT 12 backplane LAN 591. The HDLC address and frequency for the tuner
is also sent to the customer-end over the downstream telephony path and
registered at the customer end. This address is then used by the
customer-end equipment to filter the 30 Mbps downstream channel.
One advantage of the asymmetrical system is that a relatively large number
of casual users (300+) can be supported by a single multi-megabit
downstream transport, with an optimal amount of upstream capacity. The
implementation of the downstream matches the downstream of other cable
data modems in use and additionally provides superior, high capacity
upstream. Since casual users place lesser demands on the network (peak
utilization is lower than that of business), users can be concentrated on
the return channel, thus lowering head-end 32 costs.
The upstream channel in asymmetrical applications is still important due to
the nature of the acknowledge protocol of TCP/IP, where blocks of data
sent in the downstream must wait on an acknowledge message from the
receiver before subsequent data blocks are sent. If the upstream channel
is too small, the downstream channel will stall, reducing the utilization
of the downstream bandwidth. By guaranteeing a minimum of 64 Kbps to each
user, the asymmetrical system can deliver greater than 1 Mbps sustained to
each user, matching the capacity of most residential computer equipment.
Another advantage is the superior security of OFDM in the upstream. Unlike
other shared upstream modem products currently available, the asymmetrical
system herein described prevents information, such as bank accounts and
credit card numbers exchanged during on-line Internet shopping, from being
"seen" by other modems on the network
Summary of Data Delivery System Advantages
Thus, the symmetrical embodiment of system 500 provides many options for
the delivery of data services over HFC distribution network 11 to the
residence or business. The DMSM 550 provides from 64 Kbps to 512 Kbps
access to head-end resources over a 10 BaseT connection or RS232 (64 Kbps
service). The service is symmetrical (same data rate upstream and
downstream), non-shared and dedicated to each user, providing a guaranteed
level of service. As an add-in card to the HISU, the DMSM 550 provides
complete transport integration with telephony, supplying high-speed data
and two POTS lines to the residence.
The PCDM 540 provides the same data transport capabilities as the DMSM 550
in a standalone configuration, packaged in a traditional modem housing.
This implementation is ideal for premises or installations where telephony
is not deployed.
The DMCU 560 is an MISU channel unit that provides higher data rates than
either the DMSM or PCDM-512 K. The DMCU 560 router manages four
subscribers who share up to 8.192 Mbps of symmetrical bandwidth. The
router implementation guarantees that all four subscribers on the DMCU 560
have private connections. The DMCU 560 works well for multiple dwelling
installations for Internet access and small business connections where
symmetrical, non-shared data access is required.
At the head end, the LANU 580 provides the concentration of up to 100 DS0s
in flexible combinations of various data rates, from 641 Kbps to 512 Kbps
for residential, and up to 8.192 Mbps for business applications on a
single, industry standard 10 BaseT connection. An HDT 12 can be configured
with up to seven LANUs, concentrating up to 700 DS0s. In addition to the
industry standard transparent bridging function, the LANU also provides
the intelligence for the dynamic, adaptive allocation of bandwidth
capacity to optimize transport during times of heavy loading. This
capability enables an HFC service provider using system 10 to mix
residential and business data services in a single 6 Mhz channel without
compromising the quality of service for business connections during peak
Internet access times. Dynamic-allocation allows the customer units to
efficiently utilize the data transport by dropping connections at times of
no traffic and re-establishing them when data is ready to send Each time a
connection is established the LANU 580 will allocate bandwidth of up to a
maximum of 512 Kbps, depending upon the network load, with a minimum of 64
Kbps. Finally, the LANU collects detailed traffic statistics that can be
used for a variety of billing methods, for instance bill by connect time.
System 500 is particularly effective in meeting the special needs and
higher expectations of business applications. Businesses tend to require a
higher level of upstream signaling in order to support applications such
as telecommuting and videoconferencing. Most cable data modem network
architectures can provide only limited upstream capacity, but ADC is able
to, offer a very high capacity upstream due to the efficiency of OFDM and
frequency agility.
Guaranteed bandwidth is of equal importance to upstream capabilities.
Businesses must have fill access to their pipeline at all times,
regardless of other traffic on the network. With system 500, once a
premium user's bandwidth has been established, it cannot be diminished,
regardless of the number of users who subsequently access the network.
The security of the data being transported is also a major concern to
businesses. Security at the transport layer (encryption and secure key
exchange) and at the network layer (filtering) is provided by current
transport technologies. System 500 also provides additional security at
the physical layer, made possible by utilizing frequency scrambling within
the OFDM transport.
The symmetrical product line is well suited for "power" Internet users who
use their PC's not only for casual Web browsing but for remote LAN access,
telecommuting, real-time audio, and possibly video teleconferencing. While
these users are demanding, they are frequently early adopters of
technology who will push the limits of Internet access and Internet
applications, making the symmetrical, nonshared, guaranteed quality of
service of the symmetrical products a requirement.
For both residence and business users, the symmetrical embodiment of system
500 provides for superior integration with telephony. By utilizing OFDM
transport in both the upstream and downstream the symmetrical system can
carry data in the same 6 Mhz channel as telephony traffic. This capability
is ideal for smaller installations and early deployment where efficient
use of spectrum is important In addition, OFDM provides a very secure data
delivery stem by implementing a point-to-multipoint bridge for data where
two customer premises units never share the same digital data stream.
The delivery of data over system 500 requires the efficient allocation of
available bandwidth and network management of system resources.
System 500 provides a completely scalable data architecture by dynamically
allocating bandwidth for data traffic through its utilization of a subset
of standard TR303/V5 call processing software. This system capability
gives HFC service providers the flexibility to tailor the configuration of
head-end resources to satisfy the diverse needs of their subscriber base.
Subscriber services can be provisioned at the head end as symmetrical
fixed, symmetrical variable, or asymmetrical services. As the subscriber
mix changes or subscribers upgrade service, head-end resources can be
re-provisioned to meet the new requirements. For example, users can be
easily reconfigured to upgrade from 64 Kbps service to 512 Kbps or even
from asymmetrical to symmetrical. For capacity planning, data bandwidth is
allocated as a number of DS0s to potential users with a single HDT
supporting up to 720 DS0s. The number of users supported is then a
function of service level (number of DS0S) and concentration ratio (number
of users per DS0).
To ensure that service providers have an effective tool to manage their
cable data networks, system 500 offers an integrated data/telephony
network management solution. Data management is based on industry standard
SNMP agents and MIBs (management information bases), which are then
combined into an integrated data/telephony network management environment.
Integration of data delivery and telephony into a single network
management system has several advantages:
(1) Symmetrical data, asymmetrical data, and telephony elements can be
managed by the same element manager.
(2) Less support staff is required.
(3) Better integration with billing.
(4) Better fault isolation.
(5) Lower Mean Time To Repair (MTTR).
Thus, system 500 provides a single, integrated system that can meet the
diverse needs of potential subscribers, from casual Internet browsers to
high-capacity business users. The integrated solution gives HFC service
providers a single point of network management that results in reduced
support costs, reduced staffing costs, and shortened time to turn-up new
services. Finally, the OFDM technology of system 500 provides data, video
and telephony services in a bandwidth-efficient system that reduces the
demands on a very valuable commodity for HFC service providers spectrum.
ATM Embodiment
The system 500 of the present invention can also be configured to carry
data from an Asynchronous Transport Mode (ATM) network As shown in FIGS.
114 and 115, system 10 or 500 of the present invention is modified to
include an ATM multiplexer/modulator 650 which can receive ATM data from
an ATM network 652 and modulate it onto the HFC network. In one preferred
embodiment, digital video data is delivered over ATM network 652,
multiplexed and modulated using multiplexer/modulator 650 onto the HFC
network in RF digital OFDM format on assigned data and/or telephony
channels between the head end and a subscriber, as for example described
above with respect to system 10 or 500. A digital set top box 654 receives
the digital video, formatted for example in 4.0 Mbps MPEG or equivalent,
and converts it to video for display on a television 656. A return path to
the HDT 12 over a telephony or data channel allows for interactive digital
video. A video server 658 and ATM switch 660, feeding the ATM
multiplexer/modulator 650, is shown in FIG. 115.
Embodiment of Control Aspects of Telecommunications System
Channel Manager
In one embodiment, communication system 10 of FIG. 1 includes channel
manager 900 of FIG. 59 to control various aspects of the dynamic
allocation of channels to ISUs 100. For example, channel manager 900
assigns each ISU 100 to a subband, allocates channels in the subband to an
ISU to complete a communication link, and monitors the channel to detect
and avoid use of corrupted channels. Channel manager 900 implements
further functions as described below to coordinate the use of the channels
in a 6 MHz transmission channel to ISUs 100.
Channel manager 900 may comprise software executed by a processor resident
in each CXMU 56 of each HDT 12. Channel manager 900 receives events from
board support software 902, IOC and modem communicators 904, ISU ranger
906, and administrator 908. Channel manager 900 also sends messages to IOC
and modem communicators 904 for allocation or reallocation of channels.
Channel manager 900 uses two types of channels to communicate control data
to the ISUs. First, channel manager 900 broadcasts control data over the
IOC channels to the ISUs. The control data on the IOC channels contains an
identification signal that indicates the ISU to receive the control data.
Further, channel manager 900 uses an ISU demand link channel, referred to
as an IDL channel, for non-time-critical transport of data between head
end 32 and an ISU when the data is of a size that would benefit from a
transmission channel with more bandwidth than the IOC. Typically, the data
rate for the IOC channel is 16 Kbps and the data rate for the IDL channel
is 64 Kbps due to the amount of data contained in each package or frame.
Typically, control signals contain four data bytes or less per frame or
package. The IDL channel is used to transmit data packages that are larger
than this. For example, the IDL channel is used to download software to an
ISU, provision a channel unit, transmit future channel unit functions, or
transmit protocols. In one embodiment, HDT 12 only implements one IDL at a
time. The IDL channel is described in more detail below.
Subband Assignment and Channel Allocation
Channel manager 900 is responsible for assigning an ISU to a subband and
for allocating payload channels for communications links to the ISU.
Appropriate selection of subband and payload channel improve the
performance of communication system 10. Channel manager 900 further
monitors the channels and reassigns subbands and reallocates channels as
necessary to maintain acceptable communications links between head end 32
and ISUs 100.
Subband assignment
Channel manager 900 selects a subband for an ISU in several circumstances:
during acquisition, when an HISU is assigned to a subband that has
insufficient payload channels to meet a request, and during an HISU IOC
timeout event. An IOC timeout event occurs when acknowledgments are not
received by channel manager 900 from an ISU within a specified time
period. With a timeout, it is assumed that the downstream communications
to the ISU are still in tact even though the upstream communications have
become corrupted due to noise or collisions. Thus, a message on the IOC to
retune to a new subband is assumed to reach the ISU despite the lack of an
acknowledgment.
In each case in which an ISU is assigned to a subband, channel manager 900
uses various criteria to select the subband for an ISU. FIG. 62 is a flow
chart that illustrates one embodiment of a method for assigning an ISU to
a subband. According to this method, channel manager 900 first selects a
subband. Channel manager 900 then determines whether addition of the ISU
to the subband would provide an acceptable load on the IOC channel. For
example, channel manager considers the number of ISUs assigned to a
subband. Further, channel manager considers the type of ISU and the likely
load that the ISU will place on the IOC channel. By considering these
factors, channel manager 900 can selectively distribute the load on the
IOC channels so as to facilitate timely communication of control data to
and from the ISU. This also allows channel manager 900 to evenly
distribute the ISUs over the available subbands such that a like number of
ISUs occupy each subband. Channel manager 900 also weighs the number of
available channels within the subband and their transmission quality as
recorded in the tables of channel manager 900. Channels with longer
low-error rate histories will be used first Channels previously marked bad
and reallocated for monitoring will be used last. Based on these criteria,
channel manager selects a subband for each ISU.
FIGS. 63, 64 and 65 are frequency spectrum diagrams that illustrate initial
assignment of HISUs and MISUs to various subbands in a 6 MHz transmission
channel. These Figures show that channel manager 900 attempts to evenly
distribute the ISUs across the transmission channel. As depicted in FIG.
63, channel manager 900 begins assigning subbands at the middle of the 6
MHz transmission channel. Channel manager 900 then moves out toward the
ends of the transmission channel. For example, the first HISU is assigned
to subband number 12 and the twenty-fourth HISU is assigned to subband 0.
It is noted that more than one ISU can be assigned to a subband. As
depicted in FIG. 64, channel manager 900 initially assigns the first MISU
to subbands 0 through 12 and the next MISU to subbands 11 through 23. As
depicted in FIG. 65, when HISUs and MISUs are assigned to the same
subbands, channel manager assigns the subbands so as to evenly distribute
the ISUs over the available subbands. It is noted that the factors listed
for use in selecting a subband are shown by way of example and not by way
of limitation. Other factors can be added and the weight given to each
factor can be adjusted without departing from the spirit and scope of the
present invention.
Channel Allocation
FIG. 60 is a flow chart that illustrates one embodiment for a method for
allocating payload channels in a subband by channel manager 900. Channel
manager 900 attempts to maintain an acceptable distribution of bandwidth
within a subband to reduce the need for reallocation of payload channels
within the subband Further, the goal is to allocate channels appropriately
across the 6 MHz transmission channel to avoid having to reallocate
channels that are currently in use. A channel can be allocated to an ISU
only from the available channels in the subband to which the ISU is
assigned
Channel manager 900 receives a request for allocation of a payload channel
from either the SCNU 58 or CTSU 54. At block 912, channel manager 900
decides whether sufficient payload channels are available in the current
subband to fulfill the request. If sufficient channels are available, the
method proceeds to block 914 and determines whether one of the available
channels is the IDL channel. If the IDL channel is not one of the
available channels, channel manager 900 allocates a channel for each
channel requested by CTSU 54 or SCNU 58 at blocks 916 and 918. Channel
manager 900 selects the channels based on several criteria that increase
the likelihood of achieving a connection with acceptable quality levels.
For example, channel manager 900 can use the method shown in FIG. 61.
According to this method, channel manager 900 begins the selection process
by identifying available payload channels that are located toward the
center of the 6 MHz transmission channel. Typically, channels that are
nearer to the edge of the 6 MHz channel exhibit higher bit error rates
than the channels that are closer to the center. Further, channel manager
900 can also consider limitations of the ISU and the requested service in
selecting a payload channel. For example, the ISU may be preset for use
only with odd or even payload channels. This information may be included
in a ROM on the ISU and provided to the channel manager when channel
allocation is requested or during acquisition. Further, channel manager
900 uses data on the quality of transmissions over the identified channels
stored in tables in channel manager 900 to determine which available
payload channels have an acceptable error history, e.g., bit error rate.
Other appropriate criteria can be used in channel selection that also tend
to increase the chances of producing a connection with acceptable quality.
Based on these criteria, channel manager selects a payload channel to
allocate to the ISU.
If, at block 914, channel manager 900 determines that one of the available
channels is the IDL channel, channel manager 900 deallocates the payload
channel allocated to be the IDL channel at blocks 920 and 922 due to the
lower priority of communications over the IDL channel.
If, at block 912, channel manager 900 determines that sufficient payload
channels are not available in the current subband, channel manager 900
determines whether the request is for an HISU 68 or an MISU 66 at block
924. If the request is for an MISU 66, channel manager 900 sends a message
to the requestor that the request has failed at block 926.
If, at block 924, channel manager determines that the request is for an
HISU, then channel manager 900 selects a different subband at block 928 by
weighing the criteria as described above with respect to selecting a
subband. Channel manager 900 further marks the channels unavailable in the
new subband at block 930 and deallocates channels allocated to the ISU in
the prior subband at block 932. At block 934, channel manager 900 assigns
the new subband and proceeds to allocate channels as necessary at blocks
916 and 918.
An example of reassigning an ISU to a new subband to accommodate a request
for a payload channel is shown in FIGS. 66 and 67. In this example, ISUs
A, B, C, and D are initially assigned to subband 4 and ISUs E, F, and G
are assigned to subband 17 as depicted in FIG. 66. In subband 4, all
payload channels except payload channel 0 are allocated. In this case,
channel manager 900 receives a request for two payload channels for ISU C.
Since only one payload channel is available, channel manager 900 reassigns
ISU C to subband 17 which has sufficient payload channels available to
handle the current load of ISU C plus the additional two payload channels
as shown in FIG. 67.
Channel reallocation
Channel monitoring and allocation or reallocation based thereon may be used
to avoid ingress. External variables can adversely affect the quality of a
given channel. These variables are numerous, and can range from
electro-magnetic interference to a physical break in an optical fiber. A
physical break in an optical fiber severs the communication link and
cannot be avoided by switching channels, however, a channel which is
electrically interfered with can be avoided until the interference is
gone. After the interference is gone the channel could be used again
Channel Monitoring
Channel monitor 900 monitors the payload channels for errors to help in
determining which channels are acceptable for transmission for specific
services. One input to channel manager 900 is parity errors which are
available from hardware per the DS0+ channels; the DS0+ channels being
10-bit channels with one of the bits having a parity or data integrity bit
inserted in the channel as previously discussed. The parity error
information on a particular channel is used as raw data which is sampled
and integrated over time to arrive at a quality status for that channel.
In one embodiment, parity errors that are detected on downstream payload
channels are communicated to head end 32 over an associated upstream
channel. When the error is detected in the downstream transmission, the
parity bit for the upstream transmission is corrupted by intentionally
setting the parity bit to the wrong value to indicate the incorrect parity
in the downstream transmission path. Thus, the ISU informs the head end of
errors in the downstream path.
To monitor the payload channels, channel manager 900 needs an active
upstream signal on each payload channel. However, at any given time, some
payload channels may not be allocated and some allocated channels may not
be active. Thus, these payload channels do not provide the necessary
upstream signals to the head end to monitor the quality of the payload
channels. To compensate for these idle and unallocated payload channels,
channel manager 900 places these channels in loop back mode to monitor the
quality. In this case, channel manager 900 sets up the payload channel,
transmits data to the ISU on the payload channel and the ISU transmits
back specified data on an associated upstream payload channel. Channel
manager 900 monitors these channels at the head end to determine error
rates for the channels. Thus, the unallocated or idle payload channel can
be monitored for errors the same as with active channels. The goal of
channel manager 900 is to have payload established on all of the payload
channels at a given time. However, it may be acceptable to monitor the
performance of each channel at least once an hour if not active.
Channel manager 900 randomly selects and uses ISUs to monitor payload
channels in loopback mode described above. This provides several benefits
to the system. First, this allows channel manager to handle the diverse
layout of a cable plant. Channel manager 900 sets up and uses paths over
different legs from the various ODNs of the system. Further, random
cycling of the ISUs used in the loop back mode allows the system to
properly distribute power in the coaxial network. Specifically, this
random selection of ISUs for loopback mode applies to concentration type
services.
As described below, some ISUs are powered down when not active. When the
ISU is powered down, the upstream modem at the head end detects this
condition and sends a specified signal to the CXMC so that channel manager
does not use the ISU for loop back purposes. Thus, powered down ISUs do
not produce unnecessary errors.
FIG. 68 is a flow chart that illustrates a method for monitoring payload
channels by channel manager 900. Channel manager 900 reads parity error
registers of the CXMU 56 are read every 10 milliseconds. Generally, the
error counts are used to update the channel quality database and determine
which (if any) channels require reallocation. The database of channel
manager 900 contains an ongoing record of each channel. An accumulator
sums the errors with previously recorded errors to update the database.
The database organizes the history of the channels in categories such as:
current ISU assigned to the channel, start of monitoring, end of
monitoring, total error, errors in last day, in last week, number of
seconds since last error, severe errors in last day, in last week, and
current service type, such as ISDN, assigned to the channel. When the
channel is a regular (non-loop back) payload channel, channel manager 900
determines whether the performance statistics in the database are within
service specific threshold. When the statistics unacceptably exceed the
threshold, channel manager 900 reallocates the channel using a "make
before break" procedure to reduce the disruption from reallocating the
channel. Thus, channel manager 900 allocates the new payload channel for
the connection before deallocating the current payload channel.
Two issues presented by periodic parity monitoring as described above must
be addressed in order to estimate the bit error rate corresponding to the
observed count of parity errors in a monitoring period to determine if a
channel is corrupted. The first is the nature of parity itself. Accepted
practice for data formats using block error detection assumes that an
errored block represents one bit of error, even though the error actually
represents a large number of data bits. Due to the nature of the data
transport system, errors injected into modulated data are expected to
randomize the data. This means that the average errored frame will consist
of four (4) errored data bits (excluding the ninth bit). Since parity
detects only odd bit errors, half of all errored frames are not detected
by parity. Therefore, each parity (frame) error induced by transport
interference represents an average of 8 (data) bits of error. Second, each
monitoring parity error represents 80 frames of data (10 ms/125 .mu.s).
Since the parity error is latched, all errors will be detected, but
multiple errors will be detected as one error.
The bit error rate (BER) used as a basis for determining when to reallocate
a channel has been chosen as 10.sup.-3. Therefore, the acceptable number
of parity errors in a one second interval that do not exceed 10.sup.-3
must be determined. To establish the acceptable parity errors, the
probable number of frame errors represented by each observed (monitored)
parity error must be predicted. Given the number of monitored parity
errors, the probable number of frame errors per monitored parity error,
and the number of bit errors represented by a frame (parity) error, a
probable bit error rate can be derived.
A statistical technique is used and the following assumptions are made:
1. Errors have a Poisson distribution, and
2. If the number of monitored parity errors is small (<10) with respect
to the total number of "samples" (100), the monitored parity error rate
(MPER) reflects the mean frame error rate (FER).
Since a monitored parity error (MPE) represents 80 frames, assumption 2
implies that the number of frame errors (FEs) "behind" each parity error
is equal to 80 MPER. That is, for 100 parity samples at 10 ms per sample,
the mean number of frame errors per parity error is equal to 0.8 times the
count of MPEs in one second For example, if 3 MPEs are observed in a one
second period, the mean number of FEs for each MPE is 2.4. Multiplying the
desired bit error rate times the sample size and dividing by the bit
errors per frame error yields the equivalent number of frame errors in the
sample. The number of FEs is also equal to the product of the number of
MPE, and the number of FEs per MPE. Given the desired BER, a solution set
for the following equation can be determined.
##EQU4##
The Poisson distribution, as follows, is used to compute the probability of
a given number of FEs represented by a MPE (.chi.), and assumption 2,
above, is used to arrive at the mean number of FEs per MPE (.mu.).
##EQU5##
Since the desired bit error rate is a maximum, the Poisson equation is
applied successively with values for .chi. of 0 up to the maximum number.
The sum of these probabilities is the probability that no more than .chi.
frame errors occurred for each monitored parity error.
The result for a bit error rate of 10.sup.-3 and bit errors per frame error
of 1 and 8 are shown in Table 11.
TABLE 11
Bit Error Rate Probability
Prob-
Bit Errors Monitored Maximum Frame Average Frame ability
per Frame Parity Errors/Monitored Errors/Monitored of BER <
Error Errors Parity Error (x) Parity Error (.mu.) -10.sup.-3
8 2 4 1.6 98%
3 3 2.4 78%
4 2 3.2 38%
1 8 8 6.4 80%
9 7 7.2 56%
10 7 8.0 45%
Using this technique, a value of 4 monitored parity errors detected during
a one second integration was determined as the threshold to reallocate
service of an ISU to a new channel. This result is arrived at by assuming
a worst case of 8 bit errors per frame error, but a probability of only
38% that the bit error rate is better than 10.sup.-3. The product of the
bit errors per frame, monitored parity errors and maximum frame errors per
monitored parity error must be 64, for a bit error rate of 10.sup.-3 (64
errors in 64 k bits). Therefore, when the sampling of the parity errors in
the error timer event is four or greater, the channel allocator is
notified of a corrupted channel.
DS0 reordering
Some telecommunications services use multiple DS0s (payload channels) to
form a communication link in communication system 10. For example, ISDN
uses three DS0s to form three payload channels identified namely as B1,
B2, and D. To operate properly, the DS0s typically are assigned in a
specific sequence. Once the payload channels for the service are assigned,
the channel unit associated with the service expects to receive the
payload channels in a specific order. When one of the payload channels
becomes corrupted, channel manager 900 allocates a different DS0 channel
for the corrupted channel and the sequence of the DS0s is altered.
This problem could be avoided by reallocating all three DS0s. However, this
is a time consuming process and could cause transient disruption of the
service. As an alternative, channel manager 900 can assign an address to
the DS0s when the multiple DS0 service is initiated. This address can be
used by the channel unit to reconstruct the order of the DS0s on the fly
if one or more of the DS0s is reallocated out of sequence with the other
DS0s. For example, in the channel enable signal from CXMU 56 on the IOC
channel, a BIC state signal can be used to identify the correct order for
each DS0. Thus, channel manager 900 can allocate the DS0s in any order and
the channel unit can remap the DS0s to the correct order at the ISU. It is
noted that the DS0s must still be allocated in different time slots.
ISU Data Link (IDL) Channel
The IDL is a standard payload channel that is dynamically assigned to
transmit control data between HDT 12 and ISU 100 when the amount of data
exceeds the parameters of the lower bandwidth IOC channel. The IDL channel
can provide full duplex communication or simplex broadcast from HDT 12 to
one or more ISUs 100. Channel manager 900 dynamically allocates the IDL
channel as needed for non-time critical transport of data as described
above.
The IDL messages in both directions are variable in length. The IDL data is
transmitted over HFC distribution network 11 at a rate of 64 Kbps which is
one byte per 8 kHz frame. The IDL channel uses one of the 240 payload
channels and the protocol for transmitting IDL messages is handled by the
processor on CXMU 56. The processor uses cyclical redundancy codes (CRC)
and positive acknowledgments to manage error checking of IDL messages.
The IDL channel can be used to transmit various kinds of data. For example,
the IDL channel can be used to provide data to an ISU to configure a
payload channel for use with a specific protocol. For example, the IDL
channel can be used to down load data to configure a payload channel for
use with the LAPB protocol or any other appropriate protocol, including
proprietary protocols. Similarly, the IDL channel can be used to download
software to an ISU.
Transmission over the IDL channel has a lower priority than regular payload
transmissions. Thus, channel manager 900 deallocates an IDL channel before
completion of the data transmission when channel manager 900 receives a
request that requires use of the payload channel that is currently
allocated to the IDL.
FIG. 69 is a flow chart that illustrates an embodiment of a method for
allocating a payload channel to the ISU data link. At block 330a, channel
manager 900 receives a request for an IDL channel. At block 332a, channel
manager 900 determines whether a payload channel is available. If a
payload channel is available, channel manager 900 allocates the payload
channel to the IDL channel and the data is transmitted to the identified
ISU. If, however, a channel is not available, channel manager determines
whether one of the allocated channels is idle by checking the hook state
of a line of a channel unit. If the line is on hook, then channel manager
900 reallocates the channel to the IDL channel until the IDL transmission
is complete. If however, channel manager receives a request for a
communication link to the line of the channel unit, channel manager
interrupts the IDL channel and reallocates the payload channel to the
channel unit.
Power Down
Channel manager 900 can power down an ISU during periods of non-use to
reduce energy costs of communication system 10. To power down the ISU,
channel manager 900 must determine that all conditions for powering down
the ISU are met. For example, channel manager 900 can determine if the
lines of the channel units of the ISU provide service that can be powered
down. Such services may include, for example, analog services such as POTS
and COIN. Further, the lines must be idle. For example, channel manager
900 can determine if a line is idle based on the line's hook status or
other appropriate criteria. Channel manager 900 checks the status of the
lines each time a line goes from off-hook to on-hook. Channel manager 900
further checks the status of the lines every second to monitor the hook
status. It is noted, however, that channel manager 900 will not power down
an ISU that is the monitoring ISU for a subband.
When channel manager 900 determines that an ISU can be powered down, the
ISU's transmitter is disabled. Head end 32 detects the loss of power to
the ISU and sends an idle pattern upstream to the switch. An IOC control
message to of from the IOC will power-up the ISU. The IOC traffic to or
from the ISU indicates to the background task in charge of powering down
ISUs to check the ISU against the criteria for powering down again.
It is understood that the above description is intended to be illustrative,
and not restrictive. Many other embodiments will be apparent to those of
skill in the art upon reviewing the above description. The scope of the
invention should, therefore, be determined with reference to the appended
claims, along with the fulfill scope of equivalents to which such claims
are entitled.
Further Embodiment of Telephony Transport System
With reference to FIG. 116, a general description of a hybrid fiber/coax
communications network 1006 in accordance with the present invention shall
be described. Telephony and video information from existing telephone and
video services generally shown by trunk line 1008 is received by and
processed by head end 1010. Head end 1010 includes a plurality of host
distribution terminals (HDT) 1012 for telephony data interface and video
host distribution terminal (VHDT) 1014 for video data interface. Host
distribution terminals 1012 and VHDT 1014 include transmitters and
receivers for communicating the video and telephony information between
the video and telephony signal distribution network 1006 in accordance
with the present invention and the existing telephony and video services
as represented generally by trunk line 1008.
The video information is optically transmitted downstream via optical fiber
line 1017 to splitter 1018 which splits the optical video signals for
transmission on a plurality of optical fibers 1022 to a plurality of
optical distribution nodes 1026. The HDT 1012 transmits optical telephony
signals via optical fiber link 1020 to the optical distribution nodes
1026. The optical distribution nodes 1026 convert the optical video
signals and telephony signals for transmission as electrical outputs via a
coaxial distribution system 1007 to a plurality of remote units 1042. The
electrical downstream video and telephony signals are distributed via a
plurality of coaxial lines 1029 and coaxial taps 1034 of the coaxial
distribution system 1007.
The remote units 1042 include means for transmitting upstream electrical
data signals including telephony information from telephones 1076 and data
terminals 1073 and in addition may include means for transmitting set top
box information from set top boxes 1078. The upstream electrical data
signals are provided by a plurality of remote units 1042 to an optical
distribution node 1026 connected thereto. The optical distribution node
1026 converts the upstream electrical data signals to an upstream optical
data signal for transmission via optical fiber link 1020 to the head end
1010.
The present invention shall now be described in further detail with
reference to FIGS. 116-123. The first part of the description shall
primarily deal with downstream transmission and the second part of the
description shall primarily be with regard to upstream transmission. The
video and telephony distribution network 1006 in accordance with the
present invention, includes head end 1010 which receives video and
telephony information from video and telephony service providers via trunk
line 1008. Head end 1010 includes a plurality of host distribution
terminals 1012 and a video host distribution terminal 1014. The HDT 1012
includes a transmitters and receivers for communicating telephony
information, such as T1 , ISDN, or other data services information, to and
from telephony service providers via trunk line 1008 and the VHDT 1014
includes a transmitters and receivers for communicating video information,
such as cable TV video information and interactive data of subscribers to
and from video service providers via trunk line 1008.
The VHDT 1014 transmits downstream optical signals to a splitter 1018 via
video feeder optical fiber line 1017. The passive optical splitter 1018
effectively makes four copies of the downstream high bandwidth optical
video signals. The duplicated downstream optical video signals are
distributed to the correspondingly connected optical distribution nodes
1026. One skilled in the art will readily recognize that although four
copies of the downstream video signals are created, that any number of
copies may be made by an appropriate splitter and that the present
invention is not limited to any specific number.
The splitter 1018 is a passive means for splitting broad band optical
signals without the need to employ expensive broad band optical to
electrical conversion hardware. Optical signal splitters are commonly
known to one skilled in the art and available from numerous fiber optic
component manufacturers such as Gould, Inc. In the alternative, active
splitters may also be utilized. In addition, a cascaded chain of passive
or active splitters would further multiply the number of duplicated
optical signals for application to an additional number of optical
distribution nodes and therefore increase further the remote units
serviceable by a single head end. Such alternatives are contemplated in
accordance with the present invention as described by the accompanying
claims.
The VHDT 1014 can be located in a central office, cable TV head end, or a
remote site and broadcast up to about 112 NTSC channels. The VHDT 1014
includes a transmission system like that of a LiteAMp.TM. system available
from American Lightwave Systems, Inc., currently a subsidiary of the
assignee hereof. Video signals are transmitted optically by amplitude
modulation of a 1300 nanometer laser source at the same frequency at which
the signals are received (i.e. the optical transmission is a terahertz
optical carrier which is modulated with the RF video signals). The
downstream video transmission bandwidth is about 54-725 MHz One advantage
in using the same frequency for optical transmission of the video signal
as the frequency of the video signals when received is to provide high
bandwidth transmission with reduced conversion expense. This
same-frequency transmission approach means that the modulation downstream
requires optical to electrical conversion or proportional conversion with
a photodiode and perhaps amplification, but no frequency conversion. In
addition, there is no sample data bandwidth reduction and little loss of
resolution.
Alternative embodiments of the VHDT may employ other modulation and mixing
schemes or techniques to shift the video signals in frequency, and other
encoding methods to transmit the information in a coded format. Such
techniques and schemes for transmitting analog video data, in addition to
those transmitting digital video data., are known to one skilled in the
art and are contemplated in accordance with the spirit and scope of the
present invention as described in the accompanying claims.
Telephony information is transmitted downstream by HDT 1012 via optical
fiber link 1020 to a corresponding optical distribution node 1026. A more
detailed block diagram of one of the HDTs 1012 is shown in FIG. 117. Each
HDT 1012 includes an RF modem bank 1050 which receives telephony
information via trunk line 1008. The RF modem bank 1050 includes four RF
modem modules 1052 and a protection modem module 1054. Each RF modem
module receives telephony information, for example time division
multiplexed channel signals from a public switched telephone service, via
trunk line 1008 and the telephony information modulates an analog carrier
for transmission of the downstream optical telephony data by downstream
optical telephony transmitter 1080 of downstream telephony electrical to
optical converter 1064 to a corresponding distribution node 1026. Each RF
modem module includes a transceiver 1053 and provides a downstream
electrical telephony signal in one of four frequency bandwidths, each
bandwidth being about 6 MHz in width like that of a CATV channel. Each 6
MHz bandwidth channel transmits data at 22 Mbits/sec and can provide for
transmission of 8 T1 digital telephone signals; T1 being a conventional
telephone signal where 24 voice channels are sampled at an 8 kHz rate,
with 8 bits per sample (each 8 bit conversation sample is termed a DS0).
Each of these signals from the four RF modem modules 1052 are transmitted
via coax patch cables to a combiner 82 of downstream telephony electrical
to optical converter 1064 for mission by optical transmitter 1080.
Therefore, the spectrum for the downstream optical telephony data is four
separated 6 MHz frequency bands containing 22 Mbits/sec of data within
each 6 MHz bandwidth The four 6 MHz frequency bands, separated by a guard
band as is known to one skilled in the art, are transmitted in about the
725-800 MHz bandwidth
Any number of modulation techniques may be used for transmission of the
telephony information downstream. The transmission downstream is point to
multipoint transmission using broadcast type transmission schemes. The
modulation techniques utilized and performed by RF modem module 1052 may
include quadrature phase shift keying (QPSK), quadrature amplitude
modulation (QAM), or other modulation techniques for providing the desired
data rate. Modulation techniques, such as QPSK and QAM, are known to those
skilled in the art and the present invention contemplates the use of any
such modulation techniques for downstream broadcast transmission.
The electrical to optical converter 1064 includes two transmitters 1080 for
downstream telephony transmission to protect the telephony data
transmitted. These transmitters are conventional and relatively
inexpensive narrow band laser transmitters. One transmitter is in standby
if the other is functioning properly. Upon detection of a fault in the
operating transmitter, controller 1060 switches transmission to the
standby transmitter. In contrast, the transmitter of the VHDT 1014 is
relatively expensive as compared to the transmitters of HDT 1012 as it is
a broad band analog DFB laser transmitter. Therefore, protection of the
video information, non-essential services unlike telephony data, is left
unprotected. By splitting the telephony data transmission from the video
data transmission, protection for the telephony data alone can be
achieved. If the video data information and the telephony data were
transmitted over one optical fiber line by an expensive broad band analog
laser, economies may dictate that protection for telephony services may
not be possible. Therefore, separation of such transmission is of
importance.
As an alternative embodiment for providing transmission of optical video
and telephony signals to the optical distribution nodes 1027 from head end
1010 as shown in FIG. 120, the HDT 1012 and VHDT 1014 can utilize the same
optical transmitter and the same optical fiber line 1016. The signal then
is split by splitter 1018 and four split signals are provided to the
optical distribution nodes 1027 for distribution to the remote units 1042
by the coaxial distribution system 1007 as further discussed below.
However, as described above, the optical transmitter utilized would be
relatively expensive due to its broad band capabilities, lessening the
probabilities of being able to afford protection for essential telephony
services.
As one skilled in the art will recognize, optical link 1020, as shown in
FIG. 117, may include four fibers, two for transmission downstream from
electrical to optical converter 1064 and two for transmission upstream to
optical to electrical converter 1066. With the use of directional
couplers, the number of such fibers may be cut in half. In addition, the
number of protection transmitters and fibers utilized may vary as known to
one skilled in the art and any listed number is not limiting to the
present invention as described in the accompanying claims.
RF modem bank 1050 includes a protection RF modem module 1054 with a
transceiver 1053 connected to combiner 1082 of electrical to optical
converter 1064. Protection RF modem module 1054 is further coupled to
controller 1060. When a fault is detected with regard to the transmission
of one of the RF modem modules 1052, a signal is generated and applied to
an input 1062 of controller 1060. Controller 1060 is alerted to the fault
and provides appropriate signaling to switch the protection RF modem
module 1054 for the faulted RF modem such that the protection RF modem
module 1054 transmits within the 6 MHz bandwidth of the faulted RF modem
module 1052 so that the four 6 MHz bandwidth signal transmission is
continued on optical fiber link 1020. The use of one protection RF modem
module 1054 for four RF modem modules 1052 is only one embodiment of the
present invention and the number of protection RF modem modules relative
to RF modem modules may vary as known to one skilled in the art and
described in the accompanying claims. As shown in FIG. 123, RF modem bank
1050 may include one protection module 1054 for each RF modem module 1052.
In this embodiment, the RF modem bank 1050 includes three RF modem modules
1052 and three protection modules 1054 for one-to-one protection.
An optical distribution node 1026 as shown in FIG. 118 receives both the
downstream optical telephony signal and the split downstream optical video
signal. The downstream optical video signal is applied by the optical
fiber 1022 from splitter 1018 to a downstream video receiver 1120 of
optical distribution node 1026. The optical distribution node 1026 further
includes downstream telephony receiver 1121 for receiving the downstream
optical telephony signal on optical link 1020. The optical video receiver
1120 utilized is like that available in the LiteAMp.TM. product line
available from American Lightwave Systems, Inc. The converted signal from
video receiver 1120, proportionally converted utilizing photodiodes, is
applied to bridger amplifier 1127 along with the converted telephony
signal from downstream telephony receiver 1121. The bridging amplifier
1127 simultaneously applies four downstream electrical telephony and video
signals to diplex filters 1134. The diplex filters 1134 allow for full
duplex operation by separating the transmit and receive functions when
signals of two different frequency bandwidths are utilized for upstream
and downstream transmission. There is no frequency conversion performed at
the optical distribution nodes with respect to the video or downstream
telephony signals as the signals are passed through the optical
distribution nodes to the remote units via the coaxial distribution system
in the same frequency bandwidth as they are received.
After the optical distribution node 1026 has received downstream optical
video signals via optical link 1022 and downstream optical telephony
signals via optical link 1020 and such signals are converted to downstream
electrical video and telephony signals, the four outputs of the optical
distribution nodes 1026 are applied to four coaxial cables 1029 of coaxial
cable distribution system 1007 for transmission of the downstream
electrical video and telephony signals to the remote units 1042; such
transmission occurs in about the 725-800 MHz bandwidth for telephony
signals and about the 54-725 MHz bandwidth for the downstream electrical
video signals. Each optical distribution node 1026 provides for the
transmission over a plurality of coaxial cables 1029 and any number of
outputs is contemplated in accordance with the present invention as
described in the accompanying claims.
As shown in FIG. 116, each coaxial cable 1029 can provide a significant
number of remote units with downstream electrical video and telephony
signals through a plurality of coaxial taps 1034. Coaxial taps are
commonly known to one skilled in the art and act as passive bidirectional
pick-offs of electrical signals. Each coaxial cable 1029 may have a number
of coaxial taps connected in series. In addition, the coaxial cable
distribution system may use any number of amplifiers to extend the
distance data can be sent over the coaxial portions of the network 1006.
The downstream electrical video and telephony signals are provided from the
coaxial taps to the remote units 1042 in a number of different ways. In
one embodiment, the signal from the coaxial tap 1034 is provided to a home
integrated service unit 1070 as shown in FIG. 119. The home integrated
service unit 1070 of FIG. 119 includes a power tap 1099 coupled to a
conventional power supply and ring generator 1101. The downstream
electrical video and telephony signals are provided to a tap 1097 for
application of the signals to both diplex filter 1110 and ingress filter
1098. The downstream video signal is provided from ingress filter 1098 to
video equipment 1072 via set top box 1078. The downstream telephony signal
is applied from diplex filter 1110 to RF demodulator 1104 of RF modem
module 1102 and the demodulated signal is applied to an applicable service
interface for processing and connection to user equipment. For example,
the RF demodulated signal is processed via Plain Old Telephone Service
(POTS) service interface 1112 for output on twisted pairs 1118 to
telephone 1076 by POTS connection 1114. The other service interfaces such
as ISDN interface or a T1 interface perform their conventional functions
as are known to those skilled in the art for transmittal of such
information on outputs thereof to user equipment.
Ingress filter 1098 provides the remote unit 1042 with protection against
interference of signals applied to the video equipment 1072 as opposed to
those provided to other user equipment such as telephones or computer
terminals. Filter 1098 passes the video signals; however, it blocks those
frequencies not utilized by the video equipment. By blocking those
frequencies not used by the video equipment, stray signals are eliminated
that may interfere with the other services provided by the network to at
least the same remote unit
The set top box 1078 is an optional element in the network 1006. It may
include an additional modem for sending interactive data therefrom back to
head end 1010 at frequencies unused by the video and telephony
transmissions. Upstream transmission of such data is further discussed
below.
Depending on the modulation processing techniques utilized at the head end
1010, the RF demodulator 1104 would include circuitry capable of
demodulating the modulated signal. For example, if QPSK modulation is
utilized then the demodulator would include processing circuitry capable
of demodulating a QPSK modulated waveform as is known to one skilled in
the art
In another embodiment of providing downstream electrical video and
telephony signals from the coaxial taps 1034 to remote units 1042, as
shown in FIG. 116, a separate coaxial line form coaxial tap 1034 is
utilized to provide transmission of the signals therefrom to set top box
1078, and thus for providing the downstream video signals to video
equipment unit 1072. In such a case, a second coaxial line from coaxial
tap 1034 would be utilized to provide the downstream telephony signals to
a multiple integrated service unit (MISU) 1044 which would be much like
the home integrated service unit 1070 as described with regard to FIG. 119
except lacking an ingress filter 1098 and tap 1097. Unlike home integrated
service unit 1070, the MISU 1044 would be utilized to service several
remote units 1042 with telephony services via various service interfaces.
Whether the video and telephony signals are provided to the curb with use
of the MISU 1044 or whether the video and telephony signals are provided
directly to a home integrated service unit is strictly one of application
and either can be utilized with regard to the same or different coaxial
taps 1034 and within the same or different coaxial distribution systems
1007.
In addition, an optional network interface device (NID) 1075 is utilized in
the connection of telephone services to the remote units 1042, whether
they are homes or businesses, as is known to those skilled in the art and
as shown in FIG. 116. The NID is generally shown by block 1070
representing the home integrated service unit but is not shown in the
detail of FIG. 119. The NID performs service functions for the telephone
service provider such as looping back signals to the service provider that
reach the NID so as to indicate whether a failure has occurred somewhere
in transmission to the NID or in connections from the NID to the user
equipment when a failure is reported to the service provider.
The above description primarily involves the downstream transmission of
video and telephony information from head end 1010 to remote units 1042.
The upstream transmission of interactive data from set top boxes 1078 and
other data, for example telephony from telephones 1076, shall now be
described with reference to FIGS. 116-123. The description shall be
limited to transmission from remote units via home integrated service
units as transmission from an MISU is substantially similar and easily
ascertainable from such description. Home integrated service unit 1074
provides set top box information from set top box 1078 and telephony
information from the service interfaces 1112, including information from
telephone 1076, to the optical distribution mode 1026 connected thereto by
the same coaxial path as for the downstream communication. The set top box
signals are transmitted by a separate RF modem of the video service
provider at a relatively low frequency in the bandwidth of about 5 to 40
MHz which is unused by telephony and video services. The telephony signals
are also transmitted upstream in the 5-40 MHz bandwidth, usually from 10
MHz to 30 MHz. This 5-40 MHz bandwidth is reused in each coaxial path 1029
from each remote unit 1042 to the respectively connected optical
distribution node 1026. As such, upstream electrical telephony data
signals from the remote units are transmitted at the same reused frequency
bandwidth of 5-40 MHz on each coaxial line 1029 for input to the optical
distribution node 1026. Therefore, as shown in FIG. 118, four upstream
electrical telephony signals, each in the 5-40 MHz bandwidth, are input to
optical distribution node 1026, via the respectively connected coaxial
cables 1029.
The upstream transmission from an integrated service unit for multipoint to
point transmission utilizes time multiplexing techniques, although any of
a number of multiple access techniques known to those skilled in the art
are contemplated in accordance with the present invention. All the remote
units are designated a time slot for transmission. In such a case each
remote unit must transmit at a particular time to maintain multiple access
with the timing being supplied using data on the downstream paths. The
upstream data is transmitted on a bit-by-bit basis. With each remote unit
assigned a time slot, the RF modem 1102 of the unit knows that it will not
interfere with the others because it has determined the time delay for
each one of them and each RF modem 1102 is signaled to transmit at a
precise time. Due to the high volumes of multiplexed serial data from
several outlining remote stations and limited bandwidth for transmission,
short pulse durations are required for better resolution of the data
transmitted to the head end 1010. Although the data modulates a carrier
and is transmitted in the 5 to 40 MHz bandwidth by RF modulator 1108,
because of the limited bandwidth in the upstream direction, a pulse
shaping network at each remote unit is used to generate raised cosine
pulses for the rectangular or square wave bit-by-bit stream of data
transmitted along the coaxial cable in the coaxial network.
An optimal pulse shape for transmission in a band limited coaxial cable
network is determined by the use of Fourier calculations with a given set
of boundary conditions. Also, the Fourier calculations implement a
spectral limitation constraint for the purposes of limiting the spectral
content of the optimal pulse shape. Limiting the spectral content of the
pulse shape serves two functions. The first function is to limit the
spectral characteristics of the optimal pulse shape in order to prevent
phase dispersion at the receiving end of the transmission system. The
second benefit from the spectral limitation constraint is to allow the use
of relatively simple finite impulse response filters with a minimal number
of taps.
In one embodiment of the pulse shaping network as shown in FIG. 121, 50
nanosecond pulses from the RF modulator 1108 of RF modem 1102 are
transmitted to a pulse sequencer 1301 for uniform digitization. The output
from the pulse sequencer is then applied to a ten tapped finite impulse
response filter (FIR filter) 1302 with associated electronics 1303 to
provide the addition and subtraction necessary for the filtering process.
The output is sent to a line driver circuit for output to the coaxial
cable through diplex filter 1110. The optimal pulse waveform is a raised
cosine waveform. Using such pulse shaping techniques, overcomes the
difficulty of sending extremely short pulse duration information along a
band limited coaxial cable.
The upstream electrical telephony signals from a plurality of remote units,
including signals from the RF modems 1102 and from modems in set top boxes
1078, are transmitted to the respectively connected optical distribution
node 1026 as shown in FIG. 118 via the individual coaxial cables 1029. The
upstream electrical signals are applied to a diplex filter 1134
respectively connected to a coaxial cable 1029. One of the diplex filters
1134 passes the upstream electrical telephony signal applied thereto
through to combiner 1125 while the other diplex filters pass the upstream
electrical telephony signals applied thereto to frequency shifters 1128,
1130, and 1132. Frequency shifter 1128 shifts the upstream electrical
telephony signal into the 50-85 MHz bandwidth, frequency shifter 1130
shifts another upstream electrical telephony signal into the 100-135 MHz
bandwidth and frequency shifter 1132 shifts the other upstream electrical
telephony signal into the 150-185 MHz bandwidth. The shifted signals are
combined by combiner 1125 and provided to upstream telephony and set top
control transmitters 1123. The conventional optical transmitters 1123
transmit the upstream electrical telephony signal as an upstream optical
telephony signal to head end 1010 via fiber optic link 1020. Once again,
two transmitters are available for transmission, one in standby mode, like
that in the downstream transmission path.
The upstream optical telephony signals are received by upstream telephony
and set top box receiver 1084 of optical to electrical converter block
1066. The upstream optical telephony signals are converted, split, and all
the split electrical signals in the 5-40 MHz, 50-85 MHz, 100-135 MHz, and
150-185 MHz are frequency shifted back to the 5-40 MHz bandwidth by
frequency shifters 1086, 1088, and 1090 with the exception of the signal
already in the 5-40 MHz bandwidth which is passed through with the other
frequency shifted signals from the frequency shifters to RF switch 1094. A
combined signal in the 5-40 MHz bandwidth from combiner 1092 is provided
to the VHDT and the signal is processed for obtaining the interactive
information transmitted from set top boxes 1078. The RF switch 1094 is
controlled by controller 1060 and provides the upstream telephony signals
to the transceivers 1053 of the corresponding RF modems 1052. The upstream
telephony signals are then demodulated by RF modem modules 1052 and the
telephony data is provided to the service providers via trunk line 1008.
The RF modem modules 1052 include RF demodulator corresponding to the
modulation techniques utilized to transmit the information upstream so
such information can be recovered.
As discussed previously, the controller 1060 switches protection RF modem
module 1054 for a transmitting RF modem module 1052 in the downstream
communication when a fault is detected in that module. The controller also
provides signaling for switching the RF switch 1094 such that the
information which would have been provided to the faulted RF modem module
1052 is applied to the transceiver of the protection RF modem module 1054.
Therefore, the protection modem module 1054 is then fully within the
transmit and receive loop of the system.
As shown in FIG. 122, an alternative embodiment of the present invention
includes an optical to electrical converter 1066 wherein the received
optical upstream telephony signal is converted by receivers 1084 and the
entire upstream electrical signal in the 5-200 MHz bandwidths is applied
to the transceivers 1053 of the RF modem modules 1052. The RF modem
modules 1052 are then operated under control of controller 1060 which
assigns the RF modem module a carrier frequency to tune to for the
recovery of telephony information; the assigned frequency being a function
of the frequency shifting of the upstream signal. The electrical signal is
still separated and frequency shifted by frequency shifters 1086, 1088 and
1090 except for the signal already in the 5-40 MHz bandwidth and then
combined by combiner 1092 for application to VHDT 1014.
In this embodiment, the switching of the protection modem module 1054 into
the system is accomplished through the controller 1060. When the
controller 1060 detects and indicates a faulted modem module 1052, the
controller 1060 assigns the frequency previously assigned to the faulted
RF modem module to the protection module, thus establishing the protection
RF modem module 1054 fully within the transmit and receive loop.
In another embodiment shown in FIG. 123 including one-to-one protection for
the RF modem module, neither the RF switch used for protection switching
for the configuration of FIG. 123 nor the additional control required for
protection switching for the configuration of FIG. 122 is necessary. In
this embodiment, the same electrical signal provided to the RF modem
modules 1052 is applied to the corresponding protection module 1054, thus
only a control signal indicating which module is to be used for
transmission or reception is required for the one-to-one protection.
It is to be understood, however, that even though numerous characteristics
of the present invention have been set forth in the foregoing description,
together with details of the structure and function of the invention, the
disclosure is illustrative and changes in matters of shape, size, number,
and arrangement of the elements may be made within the principles of the
invention and to the fulfill extent indicated by the broad general meaning
of the terms in which the appending claims are expressed.
* * * * *
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